Power control with signal quality estimation for smart antenna communications systems

ABSTRACT

A method for power control with signal quality estimation for smart antenna communication systems is described. The method, for example, starts with a transmit power assignment. Receive weight vector determination is carried out with this assigned transmit power, and the new weights used. An estimate of the resulting received signal quality is obtained and used for another power adjustment.

CROSS-REFERENCE TO RELATED APPLICATION

This application is a continuation of U.S. patent application No.10/231,782, filed Aug. 28, 2002, entitled “Power Control with SignalQuality Estimation for Smart Antenna Communications Systems”, which, inturn, is a Continuation-In-Part of U.S. patent application No.09/020,049, filed on Feb. 6, 1998, entitled “Power Control with SignalQuality Estimation for Smart Antenna Communications Systems”, now U.S.Pat. No. 6,463,295, issued on Oct. 8, 2002, which, in turn, is aContinuation-In-Part of U.S. patent application No. 08/729,387, Filed onOct. 11, 1996, entitled “Adaptive Method For Channel Assignment In aCellular Communication System”, now U.S. Pat. No. 6,047,189, issued onApr. 4, 2000, the priority of each of which are hereby claimed.

FIELD OF INVENTION

This invention relates to the field of wireless communication systemsand more particularly to controlling radiated RF power level duringestablishment of a call and on an ongoing basis in a cellular wirelesssystem, such control of power using an estimate of the quality of areceived angle-modulated carrier.

BACKGROUND OF THE INVENTION

In a wireless communication system, as a general rule, it is highlydesirable that the minimum radiated radio frequency (RF) carrier powernecessary to achieve a specified quality level of communications be usedin order to conserve energy and, perhaps more importantly, in order toreduce interference with other users of a shared RF spectrum. With theincreasing use of cellular wireless communication systems comprising abase station (BS) at each cell, and remote terminals (a remote terminalalso is called a subscriber unit (SU) or a subscriber station)communicating with an assigned base station, the problem of interferencebetween stations within a given cellular area, and between neighboringcells, requires intelligent interference management in order to moreeffectively use the allocated common RF bandwidth. Such interferencemanagement is the goal of power control. As a general rule, the minimumradiated RF power required for maintaining an acceptable quality ofservice should be used.

Two types of power control are necessary: initial power control, andongoing power control. In initial power control, the goal is to initiatecommunications with the minimal level of power necessary to achieve anacceptable level of communications. Ongoing power control maintainsminimum transmitted power usage on a link as the communication systemchanges over time by new links being formed while others are beingestablished.

Initial Power Control

Several communications protocols are known for cellular systems,including, for example, the Personal Handiphone System (PHS) and theGlobal System for Mobile Communications (GSM). Both use time divisionmultiple access (TDMA) together with frequency division multiple access(FDMA) techniques. Such communications protocols all include protocolsfor call establishment, for example for a subscriber unit initiatingcommunications to a BS, or a BS initiating communications with a SU.Some of these protocols may not include initial power control. Therethus is a need in the art for an initial power control method that maybe applied to an existing communication system without adverselyimpacting communication system protocols that are in existence.

Ongoing Power Control

Ongoing power control is the control of radiated power as thecommunication environment changes after initial communications isachieved. For example, when the radiated power is increased in aparticular link between a SU and a BS in order to achieve an acceptablequality for the received signal, or for some other reason, such a changemay cause unacceptable quality changes for other stations using eitherthe same or adjacent channels. In addition, as new connections areestablished and on-going connections are disconnected, power assignmentsmight change resulting in changes (for better or worse) in the qualityof existing connections. For example, “excess quality” may result,implying that excess RF power is being used under the new conditions.Degraded quality also may be experienced, implying that some connectionsmay require greater radiated RF power. Variations in propagationcharacteristics, atmospherics, and man-made interference can also causechanges that require adjusting RF power levels. This is the goal ofongoing power control.

Spatial division multiple access (SDMA) techniques are known in whichthe same “conventional channel” (i.e., the same frequency channel in afrequency division multiple access (FDMA) system, timeslot in a timedivision multiple access (TDMA) system, code in a code division multipleaccess (CDMA) system, or timeslot and frequency in a TDMA/FDMA system)may be assigned to more than one subscriber station. This is done byusing an antenna array of several antenna elements at the base station,and on the uplink (communications from a subscriber unit to a basestation), the signal from each antenna element is weighted in amplitudeand phase by a receive weight (also called spatial demultiplexingweight), all the receive weights determining a complex valued receiveweight vector which is dependent on the receive spatial signature of theremote user. The receive spatial signature (also called the receivemanifold vector) characterizes how the base station array receivessignals from a particular subscriber unit. On the downlink(communications from the base station unit to a subscriber unit),transmission is achieved by weighting the signal to be transmitted byeach array element in amplitude and phase by a set of respectivetransmit weights (also called spatial multiplexing weights), all thetransmit weights for a particular user determining a complex-valuedtransmit weight vector which also is dependent on the spatial signatureof the remote user. When transmitting to several remote users on thesame conventional channel, the sum of weighted signals is transmitted atthe antenna arrays.

The weighting of the signals either on the uplink from each antennaelement in an array of antennas, or on the downlink to each antennaelement is called spatial processing herein. Spatial processing isuseful even when no more than one subscriber unit is assigned to anyconventional channel. Thus, the term SDMA shall be used herein toinclude both the true spatial multiplexing case of having more than oneuser per conventional channel, and the use of spatial processing withonly one user per conventional channel to mitigate adjacent channelinterference and adjacent cell interference, reduce the cellularfrequency reuse factor, etc. The term channel shall refer to acommunications link between a base station and a single remote user, sothat the term SDMA covers both a single channel per conventionalchannel, and more than one channel per conventional channel.

Methods for determining spatial receive and transmit weight vectors areknown in the art. See for example, U.S. Pat. No. 5,515,378 (issued May7, 1996) and U.S. Pat. No. 5,642,353 (issued Jun. 24, 1997) entitledSPATIAL DIVISION MULTIPLE ACCESS WIRELESS COMMUNICATION SYSTEMS, Roy,III, et al., inventors; U.S. Pat. No. 5,592,490 (issued Jan. 7, 1997)entitled SPECTRALLY EFFICIENT HIGH CAPACITY WIRELESS COMMUNICATIONSYSTEMS, Barratt, et al., inventors; U.S. patent application Ser. No.08/735,520 (filed Oct. 10, 1996), entitled SPECTRALLY EFFICIENT HIGHCAPACITY WIRELESS COMMUNICATION SYSTEMS WITH SPATIO-TEMPORAL PROCESSING,Ottersten, et al., inventors; U.S. patent application Ser. No.08/729,390 (filed Oct. 11, 1996) entitled METHOD AND APPARATUS FORDECISION DIRECTED DEMODULATION USING ANTENNA ARRAYS AND SPATIALPROCESSING, Barratt, et al., inventors (hereinafter “Our DemodulationPatent”); and U.S. patent application Ser. No. 08/948,772 (filed Oct.10, 1997) entitled METHOD AND APPARATUS FOR CALIBRATING A WIRELESSCOMMUNICATION STATION HAVING AN ANTENNA ARRAY, Parish, et al., inventors(hereinafter “Our Calibration Patent”), each of these incorporatedherein by reference in their entirety, these patents or applicationscollectively referred to herein as “Our Spatial Processing Patents”. Forexample, in systems that use time division duplexing (TDD) so thatuplink and downlink communications occurs over the same frequency (in aFDMA or a TDMA/FDMA system), a receive weight vector of receive weightsdetermined on the uplink can be used to determine the required transmitweight vector of transmit weights for communications on the downlinkfrom the base station to the same remote subscriber unit.

No practical methods of ongoing power control are known in the prior artthat are applicable to systems using SDMA techniques, in that the powercontrol methods can effectively adjust all of the SDMA system parametersrequired for minimizing the total radiated RF power while maintainingacceptable quality levels for all channels. Using SDMA introducessubstantial complexities in the RF radiated power control problembecause determining weight vectors affects power control, and viceversa. Any change in RF power on a conventional channel using SDMA willaffect the transmit and receive weight vectors assigned to users usingthe same conventional channel and any change in the spatial processingeffects the power required by existing users in order to maintain anadequate communication quality level. Prior art methods for powercontrol typically do not account for the specific aspects of SDMA, andmay cause instability in such an SDMA system, wherein an improper choiceof transmit power adversely alters the spatial multiplexing (i.e.,transmit) and demultiplexing (i.e., receive) weight vectors, causing thetransmit powers to deviate further from optimum until signal quality andnetwork capacity are both degraded.

The optimal solution of the ongoing power control problem for an SDMAsystem requires the simultaneous solution of the SDMA multiplexingweight assignment problem and the power assignment problem. This at thevery least is an involved computational task, and to date has been anintractable and overwhelming computational task. Thus there is a need inthe art for a practical near optimal method for determining spatialprocessing weight vectors and ongoing power control for an SDMA system.

The objective of ongoing power control problems for communications is tominimize the total power transmitted in the communication system whileensuring that a desired (“target”) signal to interference-plus-noiseratio (SINR) for every connection within every cell is achieved. Whenexpressed in this way, the resulting power control method is referred toas a globally optimal method. Such a globally optimal method in generalrequires communications between base stations of the system. Locallyoptimal methods are those for which optimality is satisfied within somesubset of the overall system, for example, within a particular cell.There may be practical difficulties with directly determining a globallyoptimal method when dealing with a large number of intercell andintracell connections. For example, the computation time may be too longrelative to the rate of change of connection conditions; and, it may notbe feasible or practical to gather the necessary data, such as the pathgain between every base station and every remote subscriber unit in realtime. It has been shown (Yun, L. C. M., Transport for Multimedia onWireless Networks, Doctoral Dissertation, University of California,Berkeley, Calif., 1995) for a non-SDMA system that, by incorporating theeffects of interference coupling between cells, the localized controlstrategy can be made to asymptotically converge to the globally optimalsolution. Thus there are advantages to having an ongoing power controlstrategy that uses locally optimal power control. Thus there is a needin the art for locally optimal power control methods for systems withSDMA that are “distributed,” in that no inter-base station communicationof power control information is required for operation.

Signal Quality Estimation

In order to implement power control, an objective measure of the qualityof the received signal is required. It is generally accepted that ameasure of the error in the signal is a useful objective measure ofquality. It is desirable that any such measurement of error be madewhile normal communications are taking place.

Several prior art methods exist for estimating the quality of receivedsignal. One class or prior art techniques uses a measure of the receivedsignal power as a measure of the received signal quality. An example isthe commonly used received signal strength indicator (RSSI). The problemwith such measures is that they do not distinguish between the desiredsignal and any interfering signals and/or noise. To overcome thisshortcoming, some prior art power control methods use a measure of thebit error rate (BER) or the easier to obtain frame error rate (FER). Forexample, the initial power control method used in the IS-95 CDMAstandard uses FER. FER is easier to obtain in practice than the BERbecause cyclic redundancy check (CRC) bits usually are part of a framestructure. The FER may be viewed as an approximate indication of theBER. Two main shortcomings of BER and FER as measures include:

-   -   1. It takes a long time (many frames) to accumulate a        statistically meaningful estimate of BER or FER, which may be        too slow for power control; and    -   2. The BER (or FER) may not be only a function of power, but may        also be affected by other causes of a demodulation error. For        example, residual frequency offset (even after any frequency        offset correction has been applied) may contribute to the        modulation error.

Additionally, prior-art decision-directed modulation error estimationmethods exist which have used for quality estimation an error vectorthat represents the difference between the received signal and anidealized model of the signal that should have been received. Theidealized model is generated from the detected bits by passing thedetected bits through a bits-to-symbol mapper which converts the bits tothe correct symbols, and then passing the correct symbols through apulse shaper to produce the idealized model of the signal (a referencesignal). The pulse shaper also needs to undo frequency correction andundo timing alignment. The difference between the resulting idealizedmodel of the modulated signal (with any frequency offset and timingmisalignment) and the actual received signal is used to estimate thenoise and interference present in the actual signal, and this is used asa quality estimate.

This prior-art decision based quality estimator has several undesirableproperties, some similar to the BER and FER measures:

-   -   1. a demodulation error may cause a large error in the quality        estimate by substituting an incorrect symbol in place of the        actual symbol that was meant to be transmitted;    -   2. frequency offset contributes to the modulation error;    -   3. measurement of modulation error does not directly relate to        the RF carrier strength and to the noise and interference        levels; and    -   4. estimation of the signal to interference-and-noise ratio        (SINR) from the modulation error tends to result in a high        variance (unreliable) estimator.

Note that the sensitivity to frequency offset is particularlyundesirable when the quality estimator is for transmitter power control.Increasing the transmitter power because a frequency offset error ismistaken for noise or interference error, is not only completelyineffective, but is undesirable because an unnecessary excesstransmitter power will cause increased interference with other systemusers.

Thus there is a need in the art for power control methods that use aprocess for estimating the quality of received signal which (a) is fast;(b) is substantially insensitive to frequency offset variations; and (c)leads to a measure, for example the signal to interference-and-noiseratio (SINR), that differentiates signal from interference and noise.

SUMMARY OF THE INVENTION

Thus one object of the present invention is a method and apparatus forongoing power control in a system that includes SDMA. Another object ofthe invention is a method and apparatus for estimating received signalquality (as expressed by the signal to interference and noise level(SINR)) for use in the power control method and for other applications.Another aspect of the invention is an initial power control method andapplication using the signal quality estimation method and/or apparatus.Yet another aspect of the invention is a method for combined initial andongoing power control applicable to a system that includes SDMA.

In one aspect of the present invention, a method for ongoing uplinkpower control for communications from one or more remote users to acommunication station with SDMA is disclosed that includes separatingthe joint determination of a spatial weight vector of weights forreceiving from a particular remote user and ongoing power control fromthat user's transmitting into a receive weight vector determining partand a separate transmit power adjustment part. The method starts withone part, for example power adjustment wherein an initial power controlstrategy is used for transmitting from the remote user. Transmit poweraccording to this initial strategy is assigned and transmission carriedout. Receive weight vector assignment is now carried out for the signalstransmitted to the communication station with this assigned transmitpower. The resulting new weight vector is used and may affect thequality of communication. An estimate of the quality of communication isobtained for communication using the newly determined receive weightvector. Ongoing power control adjustment is applied using the estimateof the quality of communication, leading to a new power assignment fortransmitting from the remote user. These new power assignments are usedleading to new receive weight vector determination. Thus iteratingbetween the transmit power setting and the spatial processing receiveweight vector determining parts, receive weight vectors and powercontrol are jointly determined.

In another aspect of the present invention, a method for ongoingdownlink power control for communications to one or more remote usersfrom a communication station with SDMA is disclosed that includesseparating the determination of a complete transmit weight vector ofweights for transmitting from the communication station to a particularremote user, the complete transmit weight vector comprising a set ofrelative transmit weights and a scaling to apply to the weights, into arelative transmit weight vector determining part and a separate transmitpower adjustment part. The method starts with one part, for examplepower adjustment wherein an initial power control strategy is used fortransmitting from the communication station to the remote user usingsome initial relative transmit weight vector of initial relativetransmit weights. Transmit power according to this initial strategy isassigned and transmission carried out. An estimate of the quality ofcommunication is obtained for communication using the initial transmitweight vector. Based on this, ongoing power control adjustment isapplied using the estimate of the quality of communication, leading to anew power assignment for transmitting from the communication stationleading to new receive weight vector determination. Separately, anupdated relative transmit weight vector is determined, and such anupdated relative transmit weight vector is used for transmitting. Thusthe complete transmit weight vector (which includes the set of relativetransmit weights and the power setting) is obtained by separatelydetermining the power setting and the spatial processing relativetransmit weight vector.

Another aspect of the invention is a method for ongoing power controlfor uplink communications between one or more remote transmitters (e.g.,SUs) and a receiving communication station (e.g., a BS), thecommunication station including an array of receiving antenna elementsand spatial receive processing according to a set of receive weights (aweight vector). The method includes for communicating with a particularremote transmitter setting up initial power assignment for theparticular transmitter, preferably according to the method described inthe Parent Patent. Starting with the initial power assignment, a set ofuplink weights (i.e., a receive weight vector) is determined at thecommunication station for the particular remote transmitter. This weightvector is used to determine a signal from the particular remotetransmitter, the determining from a plurality of signals received at theantenna elements. The received signal quality for the signals from theremote transmitters at the communication station are estimated, andbased on the received signal quality estimates, new power assignmentsare determined for the remote transmitters. The received signal qualitypreferably is an estimate of the SINR. The new power assignment isapplied at the remote transmitters. Preferably, the power assignment isdetermined at the communication station and the remote transmitters arecommanded to change power by the communication station. The remotetransmitters transmit with these new uplink powers, and the processes ofuplink weight vector determination and power control are now repeated.Preferably, power determination is carried out at prescribed intervals.

Another aspect of the invention is a method for ongoing power controlfor downlink communications between a transmitting communication station(e.g., a BS), and one or more remote receivers (e.g., a SU), thecommunication station including an array of transmitting antennaelements and spatial transmit processing to a particular remote receiveraccording to a set of transmit weights (i.e., a transmit weight vector).The method includes setting up initial power assignments for thecommunication station, preferably according to the method described inthe Parent Patent, and transmitting from the communication station usingthe initial power assignment and initial transmit weight vector, oneweight vector for each remote receiver. In the preferred embodiment, thecommunication station includes a set of receive apparatuses eachapparatus coupled to one of the antenna elements of the array forreceiving signals, and a receive processor for spatially processing thesignals received at the antenna elements from any remote transmitteraccording to a receive weight vector. In the preferred embodiment, eachremote receiver also includes a remote transmitter for transmitting asignal, and the transmit weight vector for transmitting from thecommunication station to a particular remote receiver are determinedfrom signals received at the communication station antenna array as aresult of transmitting of a signal by the remote transmitter at theparticular remote receiver, and in particular, the transmit weightvector is determined from the receive weight vector determined forreceiving the signal transmitted by the remote transmitter at theparticular remote receiver to the communication station. An estimate isdetermined of the quality of the signals received at each remotereceiver. Preferably, the signal quality estimates are each an estimateof the SINR at each remote receiver. Preferably, each remote, receiverperforms the SINR estimation and reports the received signal quality tothe communication station at prescribed intervals. Based on the signalquality estimates, downlink power assignments are determined forcommunicating with each remote receiver. The new assignments are used bythe communication station to transmit to the remote receivers. Thequality estimation, power assignment and transmission are then repeated.In the preferred embodiment, the same weight vector as previously usedis used if no weight vector updating has occurred, and an updatedtransmit weight vector is used if an updated weight vector is available.

Another aspect of the invention is a method for global ongoing powercontrol for a communication system, which includes a set of one or morecommunication stations. In the system, each communication stationcommunicates on the uplink with a set of one or more correspondingremote transmitters and on the downlink with a set of one or morecorresponding remote receivers co-located with the corresponding remotetransmitters. Preferably, the system is a cellular system, eachcommunication station is a base station, and each remote transmitter andco-located remote receiver is a subscriber unit communicating with itscorresponding base station. Each communication station including anarray of receiving antenna elements, a set of receive apparatuses coupleto the antenna array, with the outputs of the receive apparatusescoupled to a receive spatial processor, communication with a particularcorresponding remote transmitter being according to a set of receiveweights (i.e., a weight vector). Each communication station alsoincludes an array of transmitting antenna elements, a set of transmitapparatuses coupled to the antenna elements and a transmit spatialprocessor forming a set of signals for the transmit apparatuses,communication with a particular corresponding remote receiver beingaccording to a transmit weight vector. Power control for the overallsystem includes using the above method for ongoing power control foruplink communications and the above method for ongoing power control fordownlink communications. The power assignment steps in both the downlinkand the uplink ongoing power control methods include jointly determiningall the sets of transmit powers that minimize a weighted sum of all thetransmit powers for communications between the sets of correspondingremote transmitters and the communication stations (for uplink ongoingpower control) and between the communication stations and the sets ofcorresponding remote receivers (for downlink ongoing power control)under the constraint of maintaining an acceptable level of communicationfor each communication link between any transmitter (in a communicationstation on the downlink and in a remote transmitter on the uplink) andany corresponding receiver (in a corresponding remote receiver on thedownlink and in a communication station on the uplink). In oneembodiment, the weighted sum of transmit powers is the sum of alltransmit powers, the acceptable level of communications is a targetSINR, and the target SINR is the same for all uplink communications andis the same for all downlink communications.

In another embodiment of the method for global ongoing power control,the power assignment step in the ongoing power control for uplinkcommunications is carried out independently at each communicationstation and that communication station's set of corresponding remotetransmitters, and the power assignment step in the ongoing power controlfor downlink communications is carried out independently at eachcommunication station and that communication station's set ofcorresponding remote receivers.

One embodiment of the power assignment step in the ongoing power controlmethod for uplink communications includes periodically updating thepower transmitted from a remote transmitter to the communication stationas a function of the target signal quality for communicating to thecommunication station, the powers used in previous updates fortransmitting from the remote transmitter, and the estimates of theprevious quality of the signal received at the communications stationfrom the remote transmitter. The update period in the preferredembodiment is two frames. Preferably, the signal quality estimate is anSINR estimate and the target signal quality is a target SINR. In oneversion of the power assignment step, the function is only of the targetSINR, the most recent SINR estimate, and the most recently appliedpower. When all power and SINR quantities are in logarithmic scale, in aparticular version, the difference between the power to apply in thenext update and the power applied in the most recent update is somefunction of the difference between the most recent SINR estimate and thetarget SINR, and preferably this function is proportionality. In thepreferred embodiment, the target SINR is the same for all spatialchannels on a conventional channel.

One embodiment of the power assignment step in the ongoing power controlmethod for downlink communications includes periodically updating thepower transmitted from the communications station to a remote receiveras a function of the target signal quality for communicating to theremote receiver, the powers used in previous updates for transmittingfrom the communication station to the remote receiver, and the estimatesof the previous quality of the signal received at the remote receiverfrom the communications station. The update period in the preferredembodiment is two frames. Preferably, the signal quality estimate is anSINR estimate and the target signal quality is a target SINR. In oneversion of the power assignment step, the function is only of the targetSINR, the most recent SINR estimate, and the most recently appliedpower. When all power and SINR quantities are in logarithmic scale, in aparticular version, the difference between the power to apply in thenext update and the power applied in the most recent update is somefunction of the difference between the most recent SINR estimate and thetarget SINR, and preferably this function is proportionality. In thepreferred embodiment, the target SINR is the same for all spatialchannels on a conventional channel.

In another embodiment of the uplink power assignment step, on aconventional uplink channel, the set of powers to apply for uplinkcommunications for the spatial channels on the conventional uplinkchannel are those that minimize the weighted sum of the powers totransmit on the uplink spatial channels of the conventional uplinkchannel from the remote users to the communication station, constrainedby the requirement of maintaining an acceptable (i.e., target) qualityof communication on any particular uplink spatial channel of theconventional uplink channel. In a particular implementation, theminimization criterion is to minimize the total of the powers totransmit, and the constraint is that a predicted uplink signal qualitymeasure, preferably the predicted SINR on any particular uplink spatialchannel is at least some target SINR for that particular uplink spatialchannel, where the predicted uplink SINR for a particular spatialchannel is an expression of the particular spatial receive weight vectorfor the particular uplink spatial channel, the uplink path losses forthe particular uplink spatial channel and for other uplink spatialchannels of the conventional uplink channel, the receive spatialsignature of the remote transmitter on the particular uplink spatialchannel, the receive spatial signatures of the other remote transmitterson the conventional uplink channel, and the post-spatial processingnoise-plus-intercell interference experienced by the communicationstation on the particular uplink spatial channel. In a particularembodiment, the path loss for any spatial channel is a function of theestimated SINR and of the most recently used transmit power. Theintercell interference plus noise for any uplink spatial channel isdetermined as a function of the SINR estimate for that uplink spatialchannel, the receive weight vector and the receive spatial signaturesfor all uplink spatial channels on the conventional uplink channel, thepowers by the remote transmitters applied in the previous update of theuplink power control method for communicating on all the uplink spatialchannels of the conventional uplink channel, and the path losses for theparticular uplink spatial channel and for the other uplink spatialchannels on the conventional uplink channel. In a particularimplementation, the particular constraint for the particular uplinkspatial channel, denoted by subscript i, of a total number (denoted byd) of spatial channels on a conventional channel that a predicted uplinksignal quality measure, preferably the predicted SINR, is at least thevalue of a target signal quality, preferable a target SINR for theparticular uplink spatial channel (denoted by SINR_(target) _(i) ^(U)can be mathematically expressed as$\frac{L_{i}^{U}{{w_{i}^{U^{*}}a_{i}^{U}}}^{2}p_{i}^{U}}{{\sum\limits_{{j \neq i},{j = 1}}^{d}{L_{j}^{U}{{w_{i}^{U^{*}}a_{j}}}^{2}p_{j}^{U}}} + I_{i}^{U}} \geq {{SIN}\quad R_{{target}_{i}}^{U}}$where, for j=1, . . . , d, P_(j) ^(U) is the power for transmitting onuplink spatial channel j from the transmitting remote user to thecommunication station on uplink spatial channel j, L_(j) ^(U) is thepath loss (which might be a gain if larger than 1) on uplink spatialchannel j from the transmitting remote user to the communication onuplink spatial channel j, w_(j) ^(U) is the uplink (i.e. receive) weightvector (of weights) for receiving from the user on uplink spatialchannel j, with the receive weight vector having a Euclidean norm of 1,a_(j) ^(U) is the transmit spatial signature of the remote user onuplink spatial channel j, the uplink spatial signatures each havingEuclidean norm 1, and I_(j) ^(U) is the uplink post-spatial processingnoise-plus-intercell (i.e., out-of cell) interference experienced by thecommunication station on uplink spatial channel j. In a particularimplementation, the target SINRs are the same for all uplink spatialchannels of the conventional uplink channel. The uplink minimizationproblem in general is to find the set of positive p_(i) ^(U) such that$\sum\limits_{i = 1}^{d}{c_{i}^{U}p_{i}^{U}}$is minimized subject to the constraint of the predicted signal qualitybeing at least the target signal quality being met on every uplinkspatial channel of the conventional channel.

In yet another embodiment of embodiment of the uplink power assignmentstep, in a conventional uplink channel, the set of powers to apply foruplink communications for the spatial channels on any conventionaluplink channel are determined by setting the predicted uplink SINR ineach uplink spatial channel of the conventional uplink channel to beequal to a target SINR for that uplink spatial channel. In the preferredembodiment, the target SINR is the same for all uplink spatial channelsof the conventional uplink channel.

In another embodiment of the downlink power assignment step, in aconventional downlink channel, the set of powers to apply for downlinkcommunications for the spatial channels in the conventional downlinkchannel are those that minimize the weighted sum of the powers totransmit on the downlink spatial channels of the conventional downlinkchannel from the communication station to the remote receivers on theconventional channel, constrained by the requirement of maintaining anacceptable (target) quality of communication in any particular downlinkspatial channel of the conventional downlink channel. In a particularimplementation, the minimization criterion is to minimize the total ofthe powers to transmit on the conventional downlink channel and theconstraint is that a predicted downlink signal quality measure (themeasure preferable the predicted downlink SINR) for the remote receiveron any particular downlink spatial channel is at least some targetsignal quality, preferably a target SINR for the particular downlinkspatial channel, where the predicted downlink SINR for the particularspatial channel is an expression of the particular spatial transmitweight vector in the particular downlink spatial channel, the othertransmit weight vectors used for communication in the other downlinkspatial channels of the conventional downlink channel, the downlink pathlosses for the particular downlink spatial channel and for otherdownlink spatial channels of the conventional downlink channel, thetransmit spatial signature for transmitting to the remote receiver onthe particular downlink spatial channel, and the post-spatial processingnoise-plus-intercell interference experienced by the remote receiver onthe particular downlink spatial channel. In the particular embodiment,the path loss for any spatial channel is a function of the estimatedSINR at the remote receiver and of the most recently used transmitpower. The intercell interference plus noise for any spatial channel isdetermined as a function of the SINR estimate for the remote receiver onthe particular spatial channel, the transmit weight vectors and thetransmit spatial signatures for all downlink spatial channels in theconventional downlink channel, the powers applied in the previous updateof the power control method for communicating in all the downlinkspatial channels of the conventional downlink channel, and the pathlosses for the particular downlink spatial channel and for the otherdownlink spatial channels in the conventional downlink channel. In aparticular implementation, the particular constraint for the particulardownlink spatial channel, denoted by subscript i, of a total number(denoted by d) of downlink spatial channels in a conventional downlinkchannel that the predicted SINR at the remote receiver on the particulardownlink spatial channel is at least the value of the target SINR forthe particular downlink spatial channel (denoted by SINR_(target) _(i)^(D)) can be mathematically expressed as$\frac{L_{i}^{D}{{w_{i}^{D^{*}}a_{i}^{D}}}^{2}p_{i}^{D}}{{\sum\limits_{{j \neq i},{j = 1}}^{d}{L_{i}^{D}{{w_{j}^{D^{*}}a_{i}}}^{2}p_{j}^{D}}} + I_{i}^{D}} \geq {{SIN}\quad R_{{target}_{i}}^{D}}$where, for j=1, . . . , d, p_(j) ^(D) is the power for transmitting indownlink spatial channel j from the transmitting communication stationto its remote receiver on downlink spatial channel j, L_(j) ^(D) is thepath loss (which might be a gain if larger than 1) in downlink spatialchannel j from the transmitting communication station to the remotereceiver on downlink spatial channel j. w_(j) ^(D) is the downlink (i.e.transmit) weight vector (of weights) for transmitting to the user ondownlink spatial channel j, with the vectors each having Euclidean norm1, a_(j) ^(D) is the transmit spatial signature of the remote user ondownlink spatial channel j, the downlink spatial signature having aEuclidean norm of 1, and I_(i) ^(D) is the downlink post-spatialprocessing noise-plus-intercell (i.e., out-of cell) interferenceexperienced by the receiver on the particular downlink spatial channeli. The downlink minimization problem in general is to find the positiveset of p_(i) ^(D) such that$\sum\limits_{i = 1}^{d}{c_{i}^{D}p_{i}^{D}}$is minimized subject to the constraint being met on every downlinkspatial channel of the conventional downlink channel.

In yet another embodiment of embodiment of the downlink power assignmentstep, in a conventional downlink channel, the set of powers to apply fordownlink communications for the spatial channels in any conventionaldownlink channel are determined by setting the predicted downlink SINRin each downlink spatial channel of the conventional downlink channel tobe equal to a target SINR for that downlink spatial channel. In thepreferred embodiment, the target SINR is the same for all downlinkspatial channels of the conventional uplink channel.

The preferred embodiments of the ongoing power control method of thepresent invention and of the initial power control method of theinvention of the Parent Patent require an estimate of SINR of a receivedangle modulated signal. Another aspect of the invention is a method fordetermining a SINR estimate in a receiver for receiving an anglemodulated signal, the method for use in a power control method fortransmitting to the receiver, and for use in any other applicationsrequiring an estimate of the quality of a received angle modulatedsignal. In a first implementation, the method includes estimating themean amplitude level and the mean power level (i.e., the first andsecond moments of the amplitude) of the received baseband signal frommeasurements of the amplitude of the received signal and solving a setof simultaneous equations for the received SINR estimate.

In one particular embodiment of the first implementation, applicable fora signal modulated according to a digitally modulated scheme, the signalreceived in a receiver wherein the digitally modulated received signalis sampled in the receiver, the mean amplitude level and the mean powerlevel are determined from the received baseband signal amplitude samplevalues, the sample values substantially at the baud points of thedigital modulation scheme. In another particular embodiment of thisimplementation for a signal modulated according to a digital modulationscheme in a communications station having an array of antennas and asignal processor for spatial processing, wherein the digitally modulatedreceived signal is sampled in the communication station, the meanmagnitude and mean power levels are determined from the receivedbaseband signal amplitude sample values after spatial processing, thesample values substantially at the baud points of the digital modulationscheme.

Denoting the estimate magnitude of the baseband signal as R and theestimation operation as E{ }, the set of equations is${{E\lbrack R\rbrack} = {\sqrt{2\sigma^{2}}{f\left( {{SIN}\quad R} \right)}}},{where}$${{f\left( {{SIN}\quad R} \right)} = {e^{{- {SIN}}\quad R}{\sum\limits_{l = 0}^{\infty}\frac{{\Gamma\left( {\frac{3}{2} + l} \right)}{SIN}\quad R^{l}}{{\Gamma\left( {\frac{1}{2} + l} \right)}{l!}}}}},{and}$E[R²] = 2σ²(1 + SIN  R).

In one version, an iterative solution is used and values off ƒ(SINR) arepre-stored in a memory. In another version, an iterative solution alsois used and ƒ(SINR) is approximated by the value 1.

In a second implementation, the method for estimating the receivedsignal quality includes estimating the mean power level and the meanssquare of the power level (i.e., the second and fourth moments of theamplitude) of the received baseband signal from measurements of theamplitude of the received signal, and determining the SINR from theseestimates. In one particular embodiment of the second implementation,applicable for a digitally modulated signal received in a receiverwherein the digitally modulated received signal is sampled in thereceiver, the mean power level (the RSSI estimate) and the mean squaredpower level are determined from samples of the instantaneous power(i.e., the amplitude squared) substantially at the baud points. Inanother particular embodiment of this implementation for a digitallymodulated signal in a communications station having an array of antennasand a signal processor for spatial processing, wherein the digitallymodulated received signal is sampled in the communication station, themean power level and the mean squared power level are determined byaveraging post spatial processing samples of the instantaneous power(i.e., the amplitude squared) substantially at the baud points. Denotingthe RSSI (the mean power level of the received signal, post spatialprocessing in the case of SDMA) by R² , and the mean squared power by R⁴, the SINR is determined using${{{SIN}\quad R} = {\frac{\sqrt{A}}{1 - \sqrt{A}} = \frac{A + \sqrt{A}}{1 - A}}},{{{where}\quad A} = {2 - {\frac{{\overset{\_}{R}}^{4}}{\left( \overset{\_}{R^{2}} \right)^{2}}.}}}$

In the preferred embodiment of both the first and second implementation,the SINR estimate of a received signal is determined over a single timeperiod, preferably over a frame in the case of a PHS system. In animproved embodiment applicable to both the first and secondimplementation, the SINR value is determined as a running average of theSINR estimate in the current time period with SINR values determined inprevious time periods.

Other aspects of the invention will be apparent to those of ordinaryskill in the art from the following detailed description.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a block diagram of a transceiver (receiver and transmitter)module of the base station incorporating some aspects of the presentinvention. The transceiver module extracts I, Q baseband signals fromreceived RF signals for further processing in the modem module (FIG. 2),and accepts I, Q baseband signals from one or more modem modules for RFtransmission.

FIG. 2 is a block diagram of the modem module of the base stationincorporating some aspects of the present invention. The modem moduleaccepts I, Q baseband signals from one or more transceiver modules andprocesses such signals, such processing including determining signalquality and implementing power control according to various aspects ofthe invention.

FIG. 3 is a flow diagram of one embodiment of a method for callestablishment using minimal transmitted power for a PHS system;

FIG. 4 shows a phase diagram of a typical QPSK signal, includingin-phase and quadrature errors;

FIG. 5(a) shows an embodiment of an apparatus including spatialprocessing on which the signal quality aspects of the present inventionmay be realized;

FIG. 5(b) shows another embodiment of an apparatus on which the signalquality aspects of the present invention may be realized;

FIG. 6 is a flow diagram for a method for obtaining a signal qualityestimate in an angle-modulated communication systems;

FIGS. 7(a) and 7(b) each shows a flow diagram for one embodiment of acombined initial and on-going power control method. FIG. 7(a) isapplicable on the uplink and FIG. 7(b) is applicable on the downlink;

FIGS. 8(a) and 8(b) each shows a flow diagram for one embodimentapplying the on-going power control, for example, in the methods shownin the respective flow charts of FIGS. 7(a) and 7(b). FIG. 8(a) isapplicable on the uplink and FIG. 8(b) is applicable on the downlink;

FIG. 9 illustrates the RF sections of the wireless telephone subscriberunit (SU) with which the SU parts of the invention preferably areimplemented; and

FIG. 10 is a block diagram of the digital signal processor (DSP) sectionof the SU with which the SU parts of the invention preferably areimplemented.

DETAILED DESCRIPTION OF THE INVENTION

System Architecture

The methods of the invention may be implemented on any communicationsystem which includes one or more communication stations and one or moreremote receivers (for communication on the downlink) and transmitters(for communication on the uplink). The only requirement for the qualityestimation aspects of the invention is that the modulation used includessome form of angle (e.g., phase) demodulation, and the only requirementfor the downlink SDMA power control is the ability to determine downlinktransmit weights, for example from received uplink signals. The methodsof the invention preferably are implemented on a communication stationwhich is a base station and on its subscriber units, which are part of acommunication system in which a base station uses SDMA to communicatewith its subscriber units. In the preferred embodiment, thecommunication system is meant for use in a wireless local loop (WLL)cellular system. While the subscriber units in the illustrativeembodiment are fixed in location, in other systems, they may be mobile.

The SDMA base station first is described.

Base Station Architecture

A multi-element antenna array is used in the base station in order toprovide spatial processing. The number of antennas in the array isvariable. The downlink chain of the base station includes a processorfor spatially processing coupled to a set of antenna transmitapparatuses, each coupled to one of the antenna elements. The uplinkchain of the base station includes a set of antenna receive apparatuseseach receiving a signal from one of the antenna elements, with theantenna receive apparatuses coupled to a processor for spatiallyprocessing the received signals. In the illustrative embodiment,communication between base stations and subscriber units uses thestandard known as “Personal Handyphone System” (PHS), ARIB Standard,Version 2 (RCR STD-28). The PHS system is an 8-slot TDMA/FDMA systemwith true time division duplex (TDD). Each frequency channel(“subcarrier”) has an approximate bandwidth of 300 kHz. The 8 timeslotsare divided into 4 transmit (TX) timeslots and 4 receive (RX) timeslots.This implies that for any particular channel, the receive frequency isthe same as the transmit frequency. It also implies reciprocity; i.e.,the radio propagation path for both the downlink (from base station tousers' remote terminals) and the uplink (from users' remote terminals tobase station) is identical, assuming minimum motion of the subscriberunit between receive timeslots and transmit timeslots. The frequencyband of the PHS system used in the preferred embodiment is approximately1895 to 1920 MHz. Each of the 8 timeslots is 625 microseconds long. ThePHS system has a dedicated frequency and timeslot for a control channelon which call initialization takes place. Once a link is established,the call is handed to a service channel for regular communications.Communication occurs in any channel at the rate of 32 kbits per second(kbps), called full rate. Less than full rate communication is alsopossible.

In PHS as used in the preferred embodiment, a burst is defined as thefinite duration RF signal that is transmitted or received over the airduring a single timeslot. A group is defined as one set of 4 TX and 4 RXtimeslots. A group always begins with the first TX timeslot, and itstime duration is 8×0.625=5 msec.

The PHS system uses π/4 differential quaternary phase shift keying (π/4DQPSK) modulation for the baseband signal. The baud rate is 192 kbaud.That is, there are 192,000 symbols per second.

Receiver Part of the Transceiver Module in the Base Station

The base station uses an antenna array of antenna elements, and for eachantenna, a transmit/receive (T/R) switch, an analog receiver, a digitalreceiver, a digital transmitter, and an analog transmitter. Thus thismodule includes part of the antenna transmit apparatuses and part of theset of antenna receive apparatuses. The analog and digital receiver andtransmitter for any antenna element are implemented in a single RF TX/RXtransceiver module, so that each module implements a one antenna 16carrier wide band radio spanning 10 MHz of spectrum. The architecture isfairly standard, and several variations are possible, as would be clearto those of ordinary skill in the art. The particular architecture usedfor the receiver part of a transceiver module is shown in FIG. 1. The RFsignal is received at antenna element 103, and passes via a band passfilter (BPF) 104, implemented as a cavity filter with a 1895 to 1920 MHzband pass. Antenna 103 and filter 104 are external to the transceivermodule. The signal from filter 104 goes to a transmit/receive (T/R)switch 105 on the transceiver module. From switch 105, the signal passesto a low noise amplifier (LNA) 107, one or more stages of band passfiltering (not shown) and a variable attenuator 108 to a firstdownconverter 109, the downconverter using a tunable mixer using a localoscillator 111 (not part of the module) of approximately 1.6328 GHz toproduce a first IF signal 113 at 275-285 MHz (10 MHz bandwidth). Thisfirst IF signal 113 is amplified in a first IF amplifier 115 and thenpasses through a SAW BPF filter 117 (275-285 MHz) that suppress“adjacent channels” and the byproducts of the first downconverter. Theresulting signal 118 passes through a second variable attenuator 119 toa second downconverter 120 using a mixer with a tunable mixer frequencyof 291 MHz from a local oscillator 121. The output of the seconddownconverter 120 is a second IF signal 122 at −(6-16) MHz IF frequencywith a 10 MHz bandwidth and a center frequency of −11 MHz. This secondIF signal 122 is amplified in a second IF amplifier 123 and a low passfilter (LPF) 125 to an analog to digital converter (ADC) 127 whichsamples the signal at 36.864 MHz. Only the real part of the signal issampled: Thus signal 129, the output of ADC 127, contains the complexvalues IF signal centered at −11 MHz together with an image at +11 MHz,and sampling produces also an image at 25.864 MHz. This signal nowpasses through a digital downconverter/filter device 131 implementedwith an Analog Devices, Inc. (Norwood, Mass.) AD6620 Dual ChannelDecimating Receiver. In alternate implementations, a similar device, aGraychip, Inc. (Palo Alto, Calif.) GC1011 may be used, or thefunctionality may be otherwise provided for. Digitaldownconverter/filter device 131 performs several functions:

-   -   multiplying the signal by a complex phasor at a selected one of        any of the center frequencies of each of the carriers;    -   digital bandpass filtering with a desired bandpass of 300 kHz,        currently implemented as an approximately 450 kHz bandpass        filter, centered at any of the center frequencies of each of the        carriers. This gives a complex valued (in phase I part and        quadrature Q part) baseband signal three-times baud-rate        oversampled, that is, sampled 192 kHz*3=576 ksamples/sec.

The above described receiver is built on an RX/TX board, and each suchRX/TX board handles 16 received carriers, each carrier having its ownAD6620 digital downconverter/filter device. Each AD6620 device thusgenerates 16 bits of I data and 16 bits of Q data, each at 576ksamples/sec. The data is clocked out of each AD6620 device in series at18.432 MHz. This data goes to the modem board.

The Modem Module in the Base Station

A block diagram of a modem board is shown in FIG. 2. Each modem boardincludes a single general purpose processor (GPP) 203 (a Motorola/IBMPowerPC device) controlling two RX blocks and two TX blocks, each RXblock 205 including a RX data formatter 207, four RX digital signalprocessor devices (DSPs), denoted 209.1, 209.2, 209.3, and 209.4,respectively, and four RX DSP memories denoted 211.1, 211.2, 211.3, and211.4, respectively, which are connected to the four RX DSPs. There isone RX DSP and one associated RX DSP memory for each receive timeslot.Each TX block 217 includes a TX processor modulator block 221, and a TXdata formatter 225 (implemented using a FPGA). Each RX and TX block paircan handle the receive spatial processing, demodulation, modulation, andtransmit spatial processing for one carrier in a twelve antenna system,or for two carriers in a six antenna system, or three carriers in a fourantenna system. Thus one modem board can handle the necessary processingfor two carriers in a twelve-antenna system, or for four carriers in asix-antenna system, or six carriers with a four-antenna system.

A single RX block 205 of a modem board is now described. The series I, Qdata from the receive part of the transceiver of each antenna is passedto a data formatter 207 implemented as a field programmable field array(FPGA), which converts the serial stream to parallel data which isdeposited via direct memory access to one of the four RX DSP memories211.1-211.4, the data for each of the four receive timeslots going toone RX DSP memory to be processed by the associated RX DSP(209.1-209.4). The RX DSPs 209.1-209.4 are each a Motorola M56303digital signal processor. Each RX DSP performs several functions,including:

-   -   spatial processing, including determining weights;    -   frequency offset correction;    -   equalization;    -   demodulation; and    -   in one embodiment of the present invention, signal quality        estimation.

The demodulated signals output from each timeslot RX DSP 209.1-209.4 goto a signal bus, called the voice bus 231, except for certain controlsignals that go to the GPP 203 via a host port interface. The signalquality estimates determined in RX-DSPs, the SINR data sent by a remoteuser, and some status information also are sent to GPP 203. The spatialprocessing weights determined by each RX DSP go to TXprocessor/modulator block 221 in transmit block 217 via a transmitweight (TX Wt) bus 233. Transmit power adjustments as part of powercontrol are carried out by adjusting the transmit weights.

The functions performed by general-purpose processor (GPP) 203 include:

-   -   Receiving signal quality and status data from the RX DSPs;    -   In one embodiment of the present invention, using data from the        RX DSPs to determine power control; and    -   Generating all the control signals and setting all the RX DSP        and TX processor/modulator block modes and performing other        higher level functions and protocols, including communicating        with other processors in other parts of the system via an        interface FPGA 235.

Transmit block 217 operates as follows. TX processor/modulator 221accepts voice data from voice bus 231, SACCH and FACCH data from GPP 203and transmit weights for spatial processing from TX_Wt bus 233. Thefunctions of TX processor/modulator 221 include burst building,encryption, scrambling, CRC for each of the users being spatiallymultiplexed in each of the four timeslots, modulating, and applyingcomplete transmit weights (including the amplitude as power control) foreach burst just as the burst starts. Transmit block 217 can handle fourtotal spatial channels. The modulation is π/4 DQPSK, and 2* oversampledI and Q data is generated (2*192 ksamples/sec=384 ksamples/sec). Thetransmit weight applying part carries out the complex transmit weightcalculation for up to 12 transmitting antennas (i.e., a twelve elementantenna array), and for up to four spatial channels. This results in upto 12 digital signals, each having an I & Q component. The outputs of TXprocessor/modulator 221 are serialized into up to twelve differentserial data streams (I followed by Q) for each of the up to twelveantennas, each I, Q pair going to one RX/TX transceiver module. In oneimplementation, TX processor/modulator 221 includes a DSP device, memoryand a FPGA.

Transmit Part of the Transceiver Module in the Base Station

The transmit part of the transceiver module is now described with theaid of FIG. 1. Like the receiver, the transmit part can handle 16carriers of 300 kHz bandwidth for a total bandwidth of 10 MHz. Theincoming 2* upsampled baseband signal from each carrier goes to one offour Graychip, Inc. GC4114 quad digital upconverter/filter devices, eachdevice handling four carriers, for a total of four GC4114 devices on onetransceiver module. One channel of one GC4114 device is shown in FIG. 1as digital upconverter/filter 151. It performs upconversion(interpolation) of the I, Q data into a single digital signal sampled at49.152 MHz (=2*24.576 MHz), as well as adding to the present signal thesignal (carriers) from another GC4114 channel in a cascade manner, sothat the final output will be a single real valued 49.152 MHz digitalsignal of samples of a 10 MHz total bandwidth signal. This signal is fedto a 14 bit digital to analog converter (DAC) 153 to generate an analogbaseband signal 155 with 10 MHz bandwidth at center frequency −11 MHz.Signal 155 is now fed into an upconverter 157 using a mixer with atunable mixer frequency of 291 MHz from a local oscillator 121 toproduce IF signal 159 at 275-285 MHz (10 MHz bandwidth). In theillustrative embodiment, local oscillator 121 is external to thetransceiver module. Signal 159 now passes through a digitally variableattenuator 161 and then passes through an IF strip comprising two SAWfilters and two IF amplifiers, shown for simplicity in FIG. 1 as asingle BPF filter 163 and single IF amplifier 165. The filtered andamplified IF signal goes through upconverter 167, the upconvertercomprising a tunable mixer using a local oscillator 111 (external to themodule) of approximately 1.6 GHz to produce the approximately 1900 MHzRF signal 169. RF signal 169 goes through BPF 171 and then a digitallyvariable attenuator 173. This signal passes through a power amplifier(PA), a BPF and a second PA, then through a LPF. For simplicity, thiscombination of PAs and filters is shown in FIG. 1 as a single poweramplifier 175 and a single BPF 177 to produce the signal 179 that goesto T/R switch 105. The signal from switch 105 goes to the antennaelement 103 as described for the receive part.

The Subscriber Unit

FIG. 9 illustrates the RF sections of the wireless telephone subscriberunit (SU) with which the SU parts of the invention preferably areimplemented, these RF sections referred to herein by the generalreference numeral 910. RF sections 910 include a receiver front end 912and a transmitter final stage 914 each connected to an antenna 916through a bandpass filter 918 and a transmit/receive (T/R) switch 920.

The received signals go through a typical downconversion using a 1658MHz first local oscillator 922 connected to a first intermediatefrequency (IF) mixer 924 that produces an IF of 248.45 MHz. The in-phase(I) and quadrature (Q) signals are separated by an I, Q demodulator 926connected to a second local oscillator 928 operating at 469.9 MHz.

A typical local oscillator is crystal controlled and will have anaccuracy of about±10 parts per million (ppm), or±20 kHz at the 1.9 GHzRF carrier frequency. The local oscillators in the present invention arepreferably of the phase locked loop (PLL) type so that the initialcrystal frequency errors can be largely mitigated out by adjusting avoltage controlled oscillator (VCO) once the control channel isacquired. In PHS, a 20 kHz error translates to a phase error of 37.5degrees over the duration of one symbol period. It is common to usedecision-directed carrier recovery in demodulating DQPSK signals as usedin PHS. If noise is present, a decision-directed carrier recovery methodwill likely break lock, unless an initial coarse frequency correction isapplied. In the particular p/4 QPSK demodulation used in the PHSembodiment, when the frequency offset phase error reaches 45 degreesover the symbol period duration, the decision direction frequency offsetestimation will break lock completely, and the bit error rate (BER) willskyrocket.

An in-phase analog-to-digital converter (I-ADC) 30 produces 8-bitI-samples at a rate of 768 kilosamples/second. A quadrature phaseanalog-to-digital converter (Q-ADC) 32 similarly produces 8-bitQ-samples at the same rate of 768 kilosamples/second.

The transmitted signals go through a typical up-conversion using the1658 MHz local oscillator 922 connected to a final radio frequency (RF)mixer 934. The in-phase (I) and quadrature (Q) signals to be transmittedare received as a stream of 8-bit I-samples at a rate of 768kilosamples/second by an in-phase digital-to-analog converter (I-DAC)936, and as a stream of 8-bit Q-samples at the rate of 768kilosamples/second by a quadrature phase digital-to-analog converter(Q-DAC) 938.

FIG. 10 is a block diagram of a digital signal processor (DSP) section1040 that receives the I/Q-samples from the receiver front end 912 andthat produces the I/Q-signals to be sent out by the transmitter finalstage 914. The DSP section 1040 includes several DSP devices, includinga receiver-DSP (DSP(RX)) 1042 that is connected to a voice encoding DSPdevice (vocoder) DSP 1044 and a telephony interface 1046. Atransmitter-DSP (DSP(TX)) 1048 receives voice/data from the interface1046 and encodes them into the proper I/Q-signals for transmission bythe transmitter final stage 914. A fast memory 1050 supplies programexecution and support memory for the DSP(RX) 1042 and DSP(TX) 1048. AMotorola (Phoenix, Ariz.) DSP56303 24-bit digital signal processor isused for each of the DSP(RX) 1042 and DSP(TX) 1048. The DSP56303 is amember of the DSP56300 core family of programmable CMOS DSPs. Other DSPdevices or microprocessors may be substituted, as would be clear to oneof ordinary skill in the art.

Referring to FIG. 9, RF signals with carriers at approximately 1900 MHzare used to produce in-phase (“I”) and quadrature (“Q”) components thatare detected using a 469.9 MHz carrier. The I and Q signals aredigitized and sampled at four times the symbol rate. For the PHS systemused in the illustrative embodiment the symbol rate is 192 kHz, so thesampling rate in this example would be 768 kilosamples/sec. Each sampleis 8-bits deep.

In FIG. 10, the received digital I and Q signals are digital-signalprocessed by the DSP(RX) 1042. The DSP(RX) 1042 is preferably programmedto:

-   -   1. collect I and Q samples from the ADCs 1030 and 1032;    -   2. do the control-channel acquisition and processing fundamental        to time-division duplexing, do the initial estimation of        channel-control-data burst timing, and do the initial carrier        frequency offset determination;    -   3. do unpacking, frequency offset compensation, downconversion,        filtering and equalization, wherein a block of four-times        oversampled raw baseband samples corresponds to a block of        one-time oversampled (192 kHz) signals that are equalized and        baud aligned for demodulation. Time alignment to establish the        approximate baud points is carried out as described in co-owned        U.S. patent application Ser. No. 08/xxx,xxx (filed Aug. 8, 1997)        entitled METHOD AND SYSTEM FOR RAPID INITIAL CONTROL SIGNAL        DETECTION IN A WIRELESS COMMUNICATION SYSTEM, Yun, inventor. The        baud aligned I and Q samples determined by DSP(RX) 1042 are used        by DSP(RX) 1042 for estimation of the SU received signal quality        (the SU received signal to interference-plus-noise ratio) in one        aspect of the present invention;    -   4. carry out demodulation;    -   5. disassemble the demodulated burst signals;    -   6. descramble the messages;    -   7. do cyclic redundancy checks (CRC);    -   8. decrypt the traffic data;    -   9. send the voice traffic data to the vocoder DSP 1044;    -   10. send the control channel signals and channel-quality        measurements to the DSP(TX) 1048;    -   11. update the receive compensation filter and frequency offset        estimates;    -   12. in the case of SDMA, compute the calibration information to        be sent back to a base station (see, for example, Our        Calibration Patent); and    -   13. update voltage control oscillators (VCOs) and phase lock        loop (PLL) (not shown) used in the RF receiver and transmitter        part.

Thus the SU embodiments of quality estimation aspects of the presentinvention are carried out in DSP(RX) 1042.

The power for transmitting to the BS as determined according to aspectsof the present invention is adjusted using SDP(TX) 1048.

It would be clear to those in the art that the particular receiver,transmitter, and signal processing described herein for the base stationand/or for the subscriber unit is only one possible structure, and manyvariations are possible without deviating from the invention. Forexample, in the base station, the final downconversion, or upconversionneed not be carried out digitally. Similarly, the particular DSPstructure may be substituted by microprocessors or other general-purposeprocessors, or by dedicated hardware. Similarly, many othercommunication protocols may be used in place of the PHS. Finally, theinvention is not restricted to TDMA/FDMA systems.

Initial Power Control

In one aspect of the invention is a power control method thatestablishes the radiated RF power level in a communication system byadaptively controlling the transmitter power levels based on trial RFtransmissions. The communication system used in the preferred embodimentis an SDMA system meant for use in a wireless local loop (WLL) cellularsystem. One or more base stations are used , each base station (BS),also called a cell station (CS), communicating with one or moresubscriber units (SUs), a SU also called a remote terminal or personalstation (SU). Each BS includes a multi-element antenna array in order toprovide spatial processing.

In a standard PHS protocol, the control sequence for setting-up andestablishing an incoming call to a SU from the BS is shown in Table 1.TABLE 1 PHS Call Establishment Protocol 1. the BS, desiring a connectionwith a particular SU, sends a paging signal to the selected SU on acontrol channel of the selected PS, called the paging channel (PCH); 2.the selected SU responds by sending a link channel establishment requestto the BS on a control channel called the signaling control channel(SCCH); 3. the BS responds to the link channel establishment requestfrom the SU by selecting a traffic channel (TCH) and sending to the SUon the SCCH information about the selected TCH, the TCH in this casecalled the link channel (LCH); 4. the selected SU switches to theassigned LCH and transmits a sequence of synchronization (SYNCH) burstsignals followed by a sequence of idle traffic bursts; and 5. uponsuccessful detection of a synchronization signal (SYNCH burst) from theSU, the BS responds by transmitting a sequence of SYNCH bursts on theLCH followed by a sequence of idle traffic bursts, and then proceeds toestablish a connection with the incoming call to the BS, invoking anyadditional signaling that may be required (e.g., encryption and userauthentication).

The PCH is a one-way downlink point-to-multipoint channel on which theBS transmits identical information to all SUs in the paging area. TheSCCH is a bi-directional point-to-point channel that transmitsinformation needed for call connection between the BS and a SU. The TCHis a point-to-point bi-directional channel for transmitting user(subscriber) information.

It is in general desirable, and it is sometimes government policy thatthe minimal transmitter power levels adequate for each connection beused in order to reduce interference between stations using a commonfrequency band. The link establishment procedure of Table 1 does notinclude any power control, for example to ensure the use of minimaltransmitter power levels adequate for each connection, and does notaddress the impact on existing subscribers of interference that wouldresult from the new connection. Such power control is especiallycritical when bringing up a spatial channel call on a conventionalchannel already occupied by an existing user.

To satisfy the minimal transmitted power requirement, one embodiment ofthe power control method of the present invention includes having the SUintroduce a set of trial SU transmitter power levels in step 4 of theprotocol of Table 1. The initial power level used by the SU to transmita synchronization (SYNCH) burst in step 4 is set at a prescribed safelow level that generally would not be sufficient for acceptable qualityreception by the BS. Thus, the absence of a SYNCH burst reply (step 5,Table 13) indicates to the SU that a SYNCH burst was not received at theBS with acceptable quality, and thus that its SYNCH burst transmissionpower level was too low. The SU then increases the power level andretransmits a SYNCH burst, and so retransmits each time no SYNCH burstis received from the BS as would be expected in step 5. When the BSfinally receives a SYNCH burst from the SU at an acceptable quality, ittransmits a SYNCH burst to the SU, so that when a BS-transmitted SYNCHburst finally is received at the SU, the minimal transmitted power levelsufficient for communications is being used by the SU. Also, bystandardizing (a) the initial SU transmitter power level used fortransmitting the SYNCH burst, and (b) the incremental increases for eachSU retransmission, for example, +3 dB), the SU transmitter power levelrequired is determined at the BS from the number of +3 dB powerincrements that were made in the elapsed time between the link channelassignment (step 3, Table 1) and when the SU transmitted SYNCH burst wasreceived at a sufficient quality at the BS. This also can be used to setthe BS transmitted power level. The PHS system is a time-division-duplex(TDD) system, so there is substantially reciprocity of transmit andreceive propagation paths. Thus the BS can use the SU transmitter powerlevel to determine the minimum transmitter power level to be used by theBS for communicating with the SU (i.e., after taking into considerationany differences in SU and BS receiver sensitivity). For non-TDD systems,the difference in transmit and receive propagation paths may beaccounted for by performing on-air measurements and calibrating.

When a connection request originates with a SU, the connection protocolis the same as that of Table 1, but excluding step 1 of paging from theBS. Thus, modification for including power control to set minimaltransmitting power is the same as described above.

The advantage of the above described method of power control is that itmay be applied to an existing communication system without adverselyimpacting communication system protocols that are in existence.

Two embodiments of the method involve two ways of determining at the BSwhether the received signal quality is acceptable. In the firstembodiment, a measure of received signal quality is used at the BS todetermine successful reception of the SYNCH burst. The second embodimentincludes recognizing that part of a SYNCH burst that is unique to such aburst. In PHS as used in the preferred embodiment, a SYNCH burst is224-bit long and includes a 62-bit preamble and a 32-bit “Unique Word”sequence, both of which are prearranged, as well as a BS identificationcode and a SU identification code. Thus, the BS may determine successfulreception by correctly recognizing a SYNCH burst. This can be done inaddition to, or instead of using a measure of signal quality.

Table 2 shows the specification of a standard 224-bit duration SYNCHburst as used in the preferred embodiment for uplink (SU to BS) ordownlink (BS to SU) synchronization. Each of the patterns is shown inorder. TABLE 2 Name Length Description R  4 bits any 4 bit pattern SS  2bits fixed field 10 PR 62 bits a fixed periodic preamble for both uplinkand down link 0110011001100110 . . . 011001 UW 32 bits Unique Word,which for designating uplink synchronization is01101011100010011001101011110000 and for downlink synchronization is01010000111011110010100110010011; CI  4 bits fixed field 1001 CSID 42bits BS identification code PSID 28 bits SU identification code IDL 34bits all zeros, idle bits 0 . . . 00 CRC 16 bits cyclic redundancy codeerror detection.

Those of ordinary skill in the art will understand that othersynchronization signals may be used.

FIG. 3 is a flow chart that summarizes a preferred embodiment initialpower control method 301 for adaptively determining the adequate powerlevel for acceptable communications. The method of the flow chart ofFIG. 3 is designed to be compatible with the connection protocol forPHS, and does not require any modification to the PHS standard otherthan simple additions that provide downward compatibility.

Referring to FIG. 3, the method of flow chart 301 for adaptive powercontrol is presented in two versions depending on whether or not the BSis an originator of a connection request. This is shown by decision 303,which checks if the BS is the originator. If so, the method starts withstep 305 in which the BS pages the selected SU on PCH and then moves tostep 307. If the BS is not the originator, the method starts at step307. In the remainder of the description of the flow chart 301, it willbe understood that “selected SU” means the SU paged by the BS in thecase of the BS initiating the connection, and the initiating SU in thecase of a SU initiating the connection. In step 307, the selected SUsends a link channel establishment request (LCR) message to the BS onSCCH in response to the page (or, when SU originates, the originating SUsends a LCR message to the BS on SCCH). The BS selects the bestcandidate link channel (LCH) from the set of traffic channels availableand transmit the selection to the SU on SCCH as a tentatively assignedLCH in step 309. See co-owned U.S. patent application Ser. No.08/777,598 (filed Dec. 31, 1996) entitled CHANNEL ASSIGNMENT AND CALLADMISSION CONTROL FOR SPATIAL DIVISION MULTIPLE ACCESS COMMUNICATIONSYSTEMS, Yun and Ottersten, inventors, for details. At this juncture,the selected SU, at step 321 sends a SYNCH burst on the tentativelyassigned LCH at a prescribed low power level that is approximately atthe lowest possible power level at which acceptable quality reception bythe BS might be expected. At step 323, the selected SU checks if a SYNCHburst has been returned by the BS indicating that last SYNCH burst sentby the SU was received at the BS and the BS transmitted a SYNCH burst inresponse, this in turn indicating that the SU transmitted withsufficient power to establish acceptable quality reception at the BS. Ifat step 323 a BS sent SYNCH burst was not received, the SU incrementsthe transmitter power level by a prescribed amount (typically +3 dB) instep 325 and returns to step 321 to again transmit a SYNCH burst. The 3dB power increments ensure that the power level established in step 325will be within 3 dB of the minimum power required for quality reception.Finer increments would allow the established power level to be as closeas desired to the minimum power level (e.g., +1 dB increments wouldensure that the established power level is within 26% of the minimum).Meanwhile, at the BS, in step 311, the BS listens on the tentative LCHfor the SU originated SYNCH burst transmission, and in step 313 computesthe received signal quality as a signal-to-interference-plus-noise-ratio(SINR). In lieu of the SINR, one can check if the burst is correct,since all the SYNCH burst bits are known a priori. In test step 315 theBS determines if the SYNCH burst is received with acceptable quality. Ifnot, the BS waits for the next SYNCH burst from the SU. After receivinga SYNCH burst with an acceptable SINR as determined by test step 315,the BS computes (in step 317) the BS transmitting power level, thedetermination based on the time elapsed between the BS LCH assignment instep 309 and the receipt of an acceptable quality SYNCH burst in step315. Because the repeated transmissions of the SU SYNCH burst occur atprescribed intervals, 5 ms in the preferred embodiment, the power usedby the SU transmitter for the received SYNCH burst may be determined,and for the +3 dB increments of the preferred embodiment, is 2^(M−)P₀where M is the number of power increments and P₀ is the prescribedinitial SU transmitter power in linear scale (e.g., in Watts). At step319, the BS transmits a SYNCH burst using a power level based on thecomputations of step 317. In step 323, the selected SU, upon receivingthe BS SYNCH burst, recognizes that the last power level used isadequate for establishing a connection and the process ends.

Because the PHS SYNCH burst is a long bit string, the successfulreception of the SYNCH burst can be used as an optional indication thatthe received signal quality is acceptable and that the transmitter powerlevel used by the SU to transmit the received SYNCH burst is adequate.If this option is selected, the computation of the received uplink SINRin step 313 of FIG. 3 may be omitted and the test in step 315 isanswered affirmatively if the SYNCH burst bit pattern is correctlyreceived.

It should be recognized that for the purpose of clarity in describingthe method shown in FIG. 3, specific characteristics of the PHS systemhave been used. However, as previously stated, the method described isapplicable to other cellular systems and the applicability would beapparent to those practicing the art. For example, the method can beapplied to cellular systems that use Global System for MobileCommunications (GSM). GSM is very popular throughout the world, andexists also as a high frequency version called DCS-1800 and in the USAas the PCS-1900 standard for personal communication systems (PCS).Because the steps described herein for determining transmitter powerlevel are independent of the communication protocols, the entire methodcan be applied to the GSM cellular system substantially withoutmodification.

Ongoing Power Control

The preceding description of power control methods has been primarilydirected toward establishing transmitter power levels when initiating anew connection. Another aspect of the present invention is a method forcontinuing this initial power control by controlling transmitter poweron an ongoing basis in order to deal with the dynamic nature of thecommunication system, and that are applicable to SDMA systems. It willbe recognized that a communication system dynamically changes because ofthe establishment of new connections, the dropping or hand-off ofexisting connections, and changing RF propagation conditions. Thisproduces a time-varying environment in which intracell and intercellconnections interact. The establishment of new connections can causeunacceptable interference levels on existing connections, whilecanceling existing connections can reduce interference so that powerlevels remaining in use may be higher than required to maintain anacceptable quality of communication.

The goal of ongoing power control is to maximize the number of userswhile maintaining communications (as defined by some acceptable signalquality; for example, some target SINR value) for all users. For ongoingpower control, we wish to minimize the total transmit power (or, moregenerally, a weighted sum of transmit powers) while maintaining anacceptable signal quality, e.g., SINR≧SINR_(target), for all ongoingcalls. Typically, a bit error rate (BER) on the order of 10⁻³ isreasonable for voice signals encoded at 32 kbps using ADPCM,corresponding to a SINR on the order of 10 or 11 dB. In practice, toprovide a margin of safety against fading, the valueSINR_(target)≈15 dBmay be used.

For communication systems that include spatial processing techniques(SDMA techniques), including true SDMA techniques in which more than onecommunications link is possible over the same conventional channel, thecomplete ongoing power control problem can then be stated as choosingthe receive weights and the uplink transmit power (for uplink control)and the transmit weights and the transmit power (for downlink control),the downlink transmit power, for example, indicated by the relativemagnitude of the transmit weight vector.

The goal is to maximize the capacity (number of users with SINR at leastsome SINR_(target)). Note that in general, one can state this problemwith a different SINR_(target) for each spatial channel/user.

The tasks of spatial weight determining and power control are tightlycoupled. Any change in RF power on a conventional channel using SDMAwill affect the transmit and receive weights assigned to remote usersusing the same conventional channel and any change in the weights willaffect the power required by existing users in order to maintain anadequate communication quality level. The optimal solution requires thesimultaneous solution of the SDMA multiplexing weight assignment problemand the power assignment problem. For example, on the downlink, onewould determined the complete transmit weight vector, including themagnitudes, the magnitudes representing the relative transmit power onthe particular spatial channel. The simultaneous solution of the SDMAmultiplexing weight assignment problem and the power assignment problemis at the very least an involved computational task.

An aspect of the present invention is, for the uplink, to separate theuplink joint spatial multiplexing and power control problem into twoparts: a receive weight determining part and a power adjustment part.The method starts with one part, for example power control. A powercontrol strategy is used, and the transmit powers according to thisinitial strategy are assigned. Spatial receive weight assignment is nowcarried out with these assigned transmit powers. The resulting newspatial weights at first affect interference levels so that the initialpower assignment may no longer be appropriate. Using the newlydetermined spatial weights, the ongoing power control technique is againapplied, leading to new power assignments. These new power assignmentsmay mean that the receive and transmit weights are no longer optimal, sothat the new transmit powers are used as initial conditions for newtransmit and receive weights to be determined. Thus, by iteratingbetween the transmit power setting and the spatial processing weightdetermining parts, spatial weights and power control are jointlydetermined. Preferably, every new power control assignment and every newtransmit weight assignment is used immediately after the other isdetermined. Thus, the environment is constantly changing.

For downlink power control, a complete transmit weight vector can bethought of as a set of relative transmit weights, all scaled by aparticular scaling factor, so that determining a complete transmitweight vector simultaneously solves the problem of what relativetransmit weights to use for transmitting to a particular remote user,and how much power to transmit with, the power given by the scalingapplied to the relative transmit weights to form the complete weightvector. Another aspect of the present invention is, for the downlink, toseparate the complete transmit weight vector determination problem,which includes power control, into two parts: a relative transmit weightvector determining part and a power adjustment part which determines thescaling to apply to the relative transmit weight vector. The methodstarts with one part, for example power control. A power controlstrategy is used, and the transmit powers according to this initialstrategy are assigned. Relative transmit weight determination is nowcarried out with these assigned transmit powers. The resulting newrelative transmit weight vector at first affect interference levels sothat the initial power assignment may no longer be appropriate. Usingthe newly determined relative spatial transmit weights, the ongoingpower control technique is again applied, leading to new powerassignments.

Uplink power control is first described with the help of FIG. 7(a),which shows ongoing uplink power control method 701. Initially, in step703, some power is used by each SU. How to set up initial SU powerassignments in the preferred embodiment is described above in the“Initial Power Control” section and in the Parent Patent. Starting withthese power assignments, a set of uplink (i.e., receive) weight vectorsare determined at the base station. In the preferred embodiment, theuplink (receive) weight vectors are determined in step 704 (shown for aspatial channel i) by a method substantially as described in OurDemodulation Patent, incorporated herein by reference. Note that in thepreferred embodiment, these uplink weights are used by the BS todetermine downlink weights. Also note that in the preferred embodiment,the system uses time division duplexing (TDD) and the uplink anddownlink frequencies are identical for the same user. The uplink weightsare used to determine downlink weights according to methodssubstantially as described in above mentioned U.S. Pat. No. 5,592,490and in Our Calibration Patent (U.S. application Ser. No. 08/948,772,Oct. 10, 1997), both these incorporated herein by reference.

The choice of uplink weights affects the signal quality (e.g., the SINR)of the received BS (uplink) signals, so that new power control may needto be applied. Such ongoing power control is applied periodically at thebase station. In the preferred embodiment, ongoing power control isapplied after a pre-specified time period has elapsed, and this timeperiod preferably is two frames in this embodiment. Thus, in step 705for a particular spatial channel i, it is determined if it is time toapply the power control. If not, the base station waits until the nextperiod. If it is time to apply the power control, before such control isapplied, it is determined in step 708 if the call on the spatial channelshould be or has been be reassigned to another channel, or should or hasbeen handed over to another base station. If yes, the power control forthis call on this spatial channel is terminated, and this is shown bythe block labeled “END i CONNECTION” in FIG. 7(a). Otherwise, in step709, one determines the signal quality (SINR) of the received signal atthe base station. This signal quality (preferably SINR) is estimated inthe preferred embodiment using the methods described below in the“Signal Quality Estimation” section of this description, (e.g., by useof Equation (20) and FIGS. 5 and 6). Based on this, the new amounts bywhich to ramp the uplink powers up or down are determined in step 711for spatial channel i. See below for a description of some methods forstep 711. The power control according to step 711 is carried out bycommanding the remote subscriber unit (remote transmitter) on spatialchannel i. The SU transmits with these new uplink powers, and one nowreturns to step 704 of determining new uplink (receive) weights at theBS. If insufficient compute power precludes step 704 from beingperformed in the current burst or the current power control period, itmay be done at the next burst or the next power control period. Thiscloses the loop. Note again that the new uplink signals are used todetermine the downlink weights and thus affect downlink communication.

Downlink power control is now described with the help of FIG. 7(b),which shows a flow chart of the preferred embodiment ongoing powercontrol method 721. One starts with an initial set of relative transmitweights. In the preferred embodiment, these transmit weights initiallyare those determined from uplink weights which in turn are determinedfrom uplink signals after communications is established. The transmitweights preferably are normalized and thus are relative transmitweights. Alternate embodiments may use other methods to determineweights; e.g., one may determine directions of arrivals and performbeamforming, as is known in the art (see, for example, above-mentionedU.S. Pat. Nos. 5,515,378 and 5,642,353). The results of all such methodswill be described herein as weighting by a relative transmit weightvector, and the scaling of such a relative transmit vector is thepreferred way of applying the power control. The initial power to use onthe downlink on a particular spatial channel (i) to a particular SU(remote receiver) is determined and applied in step 723 using an initialpower control method, preferably the method of the Parent Patent. Thus,in step 723, the relative transmit weight vector of these initialrelative transmit weights for the particular SU on the spatial channel,is used to transmit at the initial transmit power level, resulting insignal received at the particular SU with some downlink signal quality(e.g., SINR). Estimates of these downlink SINRs are determined at theSUs, preferably using the method described below, and periodically sentto the BS. In the preferred embodiment, each SU performs the estimationand sends its signal quality estimate to the base station every frame.Other embodiments may do so as different intervals. The ongoing downlinkpower control method, like the uplink method, is applied periodically atthe base station. In the preferred embodiment, this period is every twoframes, and other embodiments may use other update periods. Thus, instep 725 for a particular spatial channel i, it is determined if it istime to apply the power control, and if not, the base station waitsuntil the next period. If it is time to apply the power control, beforesuch control is applied, it is determined in step 728 if the call onspatial channel i should be or has been reassigned to another channel,or should or has been handed over to another base station. If yes, thepower control for this call on this channel is terminated, and this isshown by the block labeled “END i CONNECTION” in FIG. 7(b). Otherwise,the downlink signal quality estimate received from the particular SU isobtained (step 729) for use by the method and this signal qualityestimate is used in step 733 for downlink power control. See below for adescription of some methods for step 733. The power control according tostep 733 is carried out by modifying the relative transmit weights todetermine actual transmit weights to use, the modification modifying themagnitudes of the relative transmit weights for any particular SU; i.e.modifying the norm of the vector of relative transmit weights. Therelative transmit weights may be the same weights as previously used ifno weight updating has occurred, or updated relative weights of ifupdated weights are available. Thus, step 731 in the flow chart showsobtaining such (new)SDMA relative transmit weights prior to applying theongoing power control step 733. As would be clear to one of ordinaryskill in the art, step 731 of obtaining updated values of the relativetransmit weight vector may occur in other points in the flow chart, andstep 733 preferably uses the latest update of the relative transmitweights.

While uplink and downlink control were described above as beingseparate, in the preferred embodiment where uplink signals and/orweights are used to determine downlink weights, uplink and downlinkcontrol is not quite partitioned. In downlink control, the initialdownlink weights and powers are based on those determined in uplinkpower control and weight determination.

The details of the alternate embodiments of determining and applying thepower control methods (steps 711 and 733 for the uplink and downlink,respectively) are now described. Several methods for carrying out apower assignment step such as steps 711 for the uplink and step 733 forthe downlink are known for conventional cellular systems, and many ofthese methods may be easily adapted for use in implementing the presentinvention. There are advantages, however, in using the novel methodssuggested herein below for the power assignment steps 711 and 733. Theinvention, however is not restricted to only using the below describedmethods for steps 711 and 733.

The Global Problem

The overall problem solved herein of what power to apply in steps 711and 733 first is described mathematically. Define p_(i) as the transmitpower for the ith transmitter (in a SU for uplink or in a BS fordownlink) in the communication system. Unless otherwise noted, all powerquantities are in a natural scale (e.g., power measurements are inWatts, not dB). The task is to determine (separately on the uplink andon the downlink) the powers pi (positive) for all i (i.e., alltransmitters) that minimize for the whole system (on the uplink, or onthe downlink) the total power. An even more general formulation is todetermine the powers p_(i) (>0) that minimize for the whole system theweighted sum of all the powers, i.e., the objective function$\begin{matrix}{{J = {\sum\limits_{i}{c_{i}p_{i}}}},} & (1)\end{matrix}$where subscript i indicates the ith transmitter, whether uplink (a SU)or downlink (a base station), p_(i) is the transmit power for the ithtransmitter, and c_(i) is a positive parameter indicating the relativeweight of the transmit power for the ith transmitter. When c_(i)=1 forall i, the criterion is to find the powers that minimize the totalpower. Note that interfering users may be intercell or intracell usersor both. For uplink determination (in step 711), there is an index i forevery uplink connection in the global system. Similarly, for downlinkdetermination (in step 733), there is an index i for every downlinkconnection in the global system. The general formulation (of havingdifferent c_(i) values) allows one to specify which connections are moreimportant than others. For example, a particular c_(i)=0 means that onthat spatial channel, no attempt to minimize the transmit power is madeso that the highest quality is maintained.

The above minimization problem is constrained by the requirement tomaintain an acceptable quality of communication. That is, the predictedSINR needs to be at least the value of the target SINR for allcommunication links. To express this mathematically, consider first theuplink. Define G_(ij) to be path loss (and/or gain) for the path fromtransmitter j to receiver i. On the uplink, the transmitter is a SU andthe receiver is a BS, while on the downlink, the transmitter is the BSand the receiver is the SU. G_(ij) includes the RF path loss experiencedbetween transmitter j and receiver i, the spatial processing rejectionor gain factor, and any other attenuation or gain factor along the pathfrom transmitter j to receiver i. Also define σ_(i) ² as the effectivebackground noise level experienced by the ith subscriber unit (on thedownlink) or BS (on the uplink) (after reception and any spatialprocessing), and define SINR_(i) as the SINR target for receiver i.

The goal then to determine, on the uplink for SU power control step 711,and on the downlink for BS power control step 733, the powers p_(i)(p_(i)>0) that minimize for the whole system the objective function ofEquation (1) while ensuring that $\begin{matrix}{\frac{G_{ii}p_{i}}{{\sum\limits_{j \neq i}{G_{ij}p_{j}}} + \sigma_{i}^{2}} \geq {{SIN}\quad{R_{{target}_{i}}.}}} & (2)\end{matrix}$

The optimization problem of Equations (1) and (2) for non-negativetransmit powers may be recognized as linear programming optimizationproblems in non-negative variables (the transmit powers). Many methodsare known for solving such linear programming problems. One well-knownmethod that may be used is the Simplex method described, for example, inMurty, K. G., Linear Programming, Wiley & Sons, New York, 1983.

Distributed Solutions

The above globally optimal method in general requires communications ofpower control information between base stations of the system. There maybe practical difficulties with directly determining a globally optimalmethod when dealing with a large number of intercell and intracellconnections. For example, the computation time may be too long relativeto the rate of change of connection conditions; and, it may not befeasible or practical to gather the necessary power control information,such as the path gain G_(ij) between every base station and every remotesubscriber unit in real time. In one embodiment, the global objectivemay be simplified to be true within some subset of the overall system,for example, within a particular cell of interest. In the case of thesubset being a particular cell, the objective of ongoing uplink powercontrol is to substantially minimize the total power transmitted by allsubscriber units within a cell of a communication system while ensuringthat the desired SINR for every connection to the BS within the cell issatisfied. Similarly, for ongoing downlink power control, the objectiveis to minimize the total power transmitted by the BS to all its SUswhile maintaining a target level of communications. The resulting uplinkand downlink power control methods wherein the objective is simplifiedto be required within some subset of the communication system, andwherein every subset is allowed to achieve its own objective, isreferred to herein as a distributed method. The distributed powercontrol strategy only requires the set of path gains between the basestation and each subscriber unit belonging to the same cell. No directbase station to base station (i.e., intercell) communication isrequired. It is possible that using only distributed, locally optimaldecisions will result in globally optimal system behavior.

The distributed method includes breaking up a global optimizationproblem into many small localized optimization problems which are solvedsimultaneously at each base station in the communication system, and foreach conventional channel on each such base station.

Method 1 for the Distributed Power Determination

The first preferred embodiments (“Method 1”) for carrying steps 711 and733 using distributed methods are now described. The procedures for boththe uplink and the downlink are at each base station to periodicallyupdate the applied power for each subscriber unit (that is, each spatialchannel), the updating based on some function of the most recently(typically the presently) applied power, the minimum acceptable signalto interference-plus-noise level (SINR), and the most recently(typically the presently) observed (i.e., estimated) SINR for thespatial channel used in communicating with the SU. This signal quality(SINR) is estimated in the preferred embodiment using the methodsdescribed below (e.g., by use of Equation (20) and FIGS. 5 and 6). Theupdating preferably every two frames, and other update periods may beused, as well as different uplink and downlink update periods. Todescribe the power determination method mathematically, let K denote themost recent (say Kth) update of the power control, let superscript idenote a particular subscriber unit, and let superscripts D and U denotedownlink (i.e., in step 733) and uplink (i.e., in step 711),respectively. Unless otherwise noted, all power and power ratios areassumed in natural (i.e., linear) units (e.g., WATTS) rather thanlogarithmic units (e.g., dB). Let p_(i)(K) be the transmitted power forthe ith user for the Kth update. Let SINR_(target) _(i) be the minimumacceptable SINR for this user, and let SINR_(i)(K) be the most recently(and usually the presently) experienced SINR for this user as determinedby an SINR estimator (estimation is indicated by the double overbar).Note the above SINR and the p_(i)(K) quantities have the D or Usuperscripts omitted for simplicity.

The power control (steps 711 on the uplink and step 733 on the downlink)then is applied as follows from update period to update period. For thenext (i.e., (K+1)th) update, the transmitter power of the ith user isupdated according to the following iterative rule. On the uplink foruser i, the updated power to use (the (K+1)'th update) is a function ofthe target SINR for that user and the previously applied powers and theprevious estimates of the SINR for that receiver. That is, for user i,$\left. {{{p_{i}^{U}\left( {K + 1} \right)} = {f^{U}\left( {\left\{ {p_{i}^{U}(J)} \right\}_{J = 1}^{K},\overset{\_}{\overset{\_}{{SIN}\quad{R_{i}^{U}(J)}}}} \right\}}_{J = 1}^{K}},{{SIN}\quad R_{{target}_{i}}^{U}}} \right),$where ƒ^(U) is some function. In one embodiment, the function ƒ^(U)includes only the most recent (i.e., Kth) and the previous to mostrecent (i.e., (K−1)'th) SINR estimate and only the most recent appliedpower. That is,${p_{i}^{U}\left( {K + 1} \right)} = {{f^{U}\left( {{p_{i}^{U}(K)},\overset{\_}{\overset{\_}{{SIN}\quad{R_{i}^{U}\left( {K - 1} \right)}}},\overset{\_}{\overset{\_}{{SIN}\quad{R_{i}^{U}(K)}}},{{SIN}\quad R_{{target}_{i}}^{U}}} \right)}.}$In a second embodiment,${p_{i}^{U}\left( {K + 1} \right)} = {{f^{U}\left( {{p_{i}^{U}(K)},\overset{\_}{\overset{\_}{{SIN}\quad{R_{i}^{U}(K)}}},{{SIN}\quad R_{{target}_{i}}^{U}}} \right)}.}$Similarly on the downlink,${{p_{i}^{D}\left( {K + 1} \right)} = {f^{D}\left( {\left\{ {p_{i}^{D}(J)} \right\}_{J = 1}^{K},\left\{ \overset{\_}{\overset{\_}{{SIN}\quad{R_{i}^{D}(J)}}} \right\}_{J = 1}^{K},{{SIN}\quad R_{{target}_{i}}^{D}}} \right)}},$where ƒ^(D) is some other function. In one embodiment, the functionƒ^(U) includes only the most recent (Kth) and the previous to mostrecent (i.e., (K−1)'th) SINR estimate and only the most recently appliedpower. That is,${p_{i}^{D}\left( {K + 1} \right)} = {{f^{D}\left( {{p_{i}^{D}(K)},\overset{\_}{\overset{\_}{{SIN}\quad{R_{i}^{D}\left( {K - 1} \right)}}},\overset{\_}{\overset{\_}{{SIN}\quad{R_{i}^{D}(K)}}},{{SIN}\quad R_{{target}_{i}}^{D}}} \right)}.}$In a second embodiment,${p_{i}^{D}\left( {K + 1} \right)} = {{f^{D}\left( {{p_{i}^{D}(K)},\overset{\_}{\overset{\_}{{SIN}\quad{R_{i}^{D}(K)}}},{{SIN}\quad R_{{target}_{i}}^{D}}} \right)}.}$

In the preferred embodiment, for all users i on the uplink step 711 anddownlink step 733, the same function is used. That is ƒ^(D)=ƒ^(U). Inparticular, $\begin{matrix}{{p_{i}^{D}\left( {K + 1} \right)} = {\left( \frac{{SIN}\quad R_{{target}_{i}}^{D}}{\overset{\_}{\overset{\_}{{SIN}\quad{R_{i}^{D}(K)}}}} \right)^{\mu}{p_{i}^{D}(K)}}} & \left( {3a} \right) \\{and} & \quad \\{{{p_{i}^{U}\left( {K + 1} \right)} = {\left( \frac{{SIN}\quad R_{{target}_{i}}^{U}}{\overset{\_}{\overset{\_}{{SIN}\quad{R_{i}^{U}(K)}}}} \right)^{\mu}{p_{i}^{U}(K)}}},} & \left( {3b} \right)\end{matrix}$respectively, where μ is some constant. The description of Equations (3)provide for different target levels of quality of communications foreach spatial channel and for the uplink and downlink. In the preferredembodiment, the target SINRs are the same for all users i and are thesame for the uplink step 711 and the downlink step 733.

This embodiment can now be described with all quantities in alogarithmic scale (e.g., power and/or SINR measurements and/or estimatesare in dB) rather than in a natural scale. Subscript L is used to denotelogarithmic scale. Again, let K denote the most recent (say Kth)iteration of power control, and let SINR_(Ltarget) _(i) be the minimumacceptable SINR (log scale) for a particular spatial channel i, and letSINR_(L) _(i) (K) be its most recently experienced SINR (log scale) asdetermined by an SINR estimator (shown independent of the SU forsimplicity). For a particular spatial channel i, let p_(Li)(K+1) be thetransmit power (in log scale) to use for the next update (two frameslater, in the preferred embodiment), and let p_(Li)(K) be the mostrecently applied power (in log scale). The power control, according tothe first preferred embodiment is to use in the next iteration the power(log scale) used in the most recent iteration plus an increment which issome function of the difference between the target SINR (log scale) andan estimate of the most recently experienced SINR (log scale).Preferably, the function is proportionality, so that the procedure is toupdate the applied power based on the most recently (e.g., thepresently) applied power and the difference between the minimumacceptable SINR and the estimated SINR (all in log scale).Mathematically, in both the uplink step 711 (with superscript U notshown but understood) and downlink step 733 711 (with superscript D notshown but understood), for each i,P _(Li)(K+1)=P _(Li)(K)+μ(SINR_(Ltarget) _(i) − SINR_(Li)(K)),  (4)where the powers and SINRs are in logarithmic scale and μ is a constant,which in the preferred embodiment is 0.12. Note that SINR_(Ltarget) _(i)may be different for each spatial channel. For example, when one hasboth voice and data traffic, for 32 kbps voice as used in PHS (adaptivedifferential PCM), a SINR_(Ltarget) of 15 dB may be good enough, whilefor a data channel, the desired BER may be in the order of 10⁻⁶ to 10⁻⁸which would correspond to an approximate SINR_(Ltarget) of 21 dB. In thepreferred embodiment, the same value, SINR_(Ltarget) is used for allspatial channels. The power control step 711 using Equation (4) withsuperscript U is carried out for each spatial channel on the uplink, andthe power control step 733 using Equation (4) with superscript D iscarried out for each spatial channel on the downlink. Note that whileEquation (4) described adjusting the power according to a linearfunction of the difference (in log scale, e.g., in dB) between thetarget and actual estimated SINR, the method in steps 711 or 733 may begeneralized to an adjustment of the transmit power (in log scale)according to some (e.g., nonlinear) function (say functioned) of thedifference between the target and actual estimated SINR:P _(Li)(K+1)=P _(Li)(K)+ƒn(SINR_(Ltarget) _(i) − SINR_(i)(K)),  (5)where the superscripts U or D are left out from each variable and ƒn forsimplicity. Again, the power control (step 711 on the uplink and step733 on the downlink) using Equation 5 is carried out for all spatialchannels at each of the cells of the communication system. No basestation to base station communication therefore need be used.

It should be noted that because this method does not require anyknowledge of the transmission gains, there is no need to model and/ormeasure the spatial processing gains, path gains, or distinguish betweenintracell vs. intercell interference. The method only requires areliable SINR estimator such as the carrier modulus moment methoddescribed below (the “signal quality estimation” aspect of theinvention).

Method 2 for the Power Determination

The second embodiment of the distributed power control problem (steps711 or 733) is to explicitly find a solution to the localizedoptimization problem for the uplink and for the downlink. Whenlocalized, the optimization problem can be stated as follows. At anysingle base station, let there be d spatial channels in the conventionalchannel. Natural (linear) scale again is used in the followingdescription for all powers and SINRs. The task is to find the positivepowers pi that minimize the weighted sum of the powers, i.e., theobjective function $\begin{matrix}{{J = {\sum\limits_{i = 1}^{d}{c_{i}p_{i}}}},} & (6)\end{matrix}$where as before subscript i indicates the spatial channel, p_(i) is thetransmit power (in linear units) for the ith spatial channel, and c_(i)is a positive parameter indicating the relative weight of the transmitpower for the ith spatial channel. In the preferred embodiment, c_(i)=1for all i so that the criterion is to minimize the total power. Inalternate embodiments, some spatial channels may be more important thanothers and different values of c_(i) may be chosen to reflect this.

The above minimization problem is constrained by the requirement tomaintain a minimum quality of communication. That is, the predicted SINRwhen one uses a particular spatial weight (say, on the downlink, w_(i)^(D)) in spatial channel (denoted i) needs to be at least the value ofthe target SINR for that spatial channel, for all spatial channels. Toexpress this mathematically, for implementation in downlink powercontrol step 733, define the following quantities:

-   -   L_(i) ^(D) is the path loss (or gain) for spatial channel i and        its associated SU;    -   w_(i) ^(D) is the downlink (i.e. transmit) multiplexing weight        vector (of weights) for user (i.e., spatial channel) i, with the        vectors each having Euclidean norm 1. That is, ∥w_(i) ^(D)∥=1        for all i where for any vector x with complex valued elements        x_(l) having real and imaginary parts x_(Rl) and x_(Il),        respectively, l=1, . . . , m,        ${{x} = \sqrt{{\sum\limits_{j = 1}^{m}x_{Rj}^{2}} + x_{Ij}^{2}}};$    -   a_(i) ^(D) is the spatial signature of the ith remote user on        the downlink on this base station, with ∥a_(i) ^(D)∥=1 for        all i. See above mentioned incorporated-by-reference U.S. Pat.        No. 5,592,490 for a formal definition of the spatial signature;        and    -   I_(i) ^(D) is the downlink post-spatial processing        noise-plus-intercell (i.e., out-of cell) interference        experienced by subscriber i.    -   |w_(i) ^(D) ^(*) a_(i) ^(D)|² is then a measure of the        beamforming gain in the direction of user i, and for j≠i, |w_(j)        ^(D) ^(*) a_(i) ^(D)|² is a measure of the gain from undesired        spatial channel j in the direction of user i, where the *        indicates the complex conjugate transpose (also called the        Hermitian transpose). The constraint that the predicted SINR        when one uses a particular spatial weight vector for spatial        channel i needs to be at least the value of the target SINR for        that spatial channel, can then be expressed as, on the downlink,        $\begin{matrix}        {{\frac{L_{i}^{D}{{w_{i}^{D^{*}}a_{i}^{D}}}^{2}p_{i}^{D}}{{\sum\limits_{{j \neq i},{j = 1}}^{d}{L_{i}^{D}{{w_{j}^{D^{*}}a_{i}^{D}}}^{2}p_{j}^{D}}} + I_{i}^{D}} \geq {{SIN}\quad R_{{target}_{i}}^{D}}},} & \left( {7a} \right)        \end{matrix}$        and the downlink optimization problem (for step 733) is to find        the positive set of p_(i) ^(D)>0 such that J $\begin{matrix}        {J^{D} = {\sum\limits_{i = 1}^{d}{c_{i}^{D}p_{i}^{D}}}} & \left( {7b} \right)        \end{matrix}$        is minimized subject to the constraint of Equation (7a), where        the quantities are as before with superscript D to indicate the        downlink.

Similarly, for use on the uplink power control step 711, usingsuperscript U to denote uplink for the same quantities as defined abovewith superscript D for the downlink, the constraint that the predictedSINR when one uses a particular spatial weight w_(i) ^(U) for spatialchannel i needs to be at least the value of the target SINR for thatspatial channel, can then be expressed as $\begin{matrix}{{\frac{L_{i}^{U}{{w_{i}^{U^{*}}a_{i}^{U}}}^{2}p_{i}^{U}}{{\sum\limits_{{j \neq i},{j = 1}}^{d}{L_{i}^{U}{{w_{i}^{U^{*}}a_{j}}}^{2}p_{j}^{U}}} + I_{i}^{U}} \geq {{SIN}\quad R_{{target}_{i}}^{U}}},} & \left( {8a} \right)\end{matrix}$and the uplink optimization problem is to find the p_(i) ^(U)>0 suchthat $\begin{matrix}{J^{U} = {\sum\limits_{i = 1}^{d}{c_{i}^{U}p_{i}^{U}}}} & \left( {8b} \right)\end{matrix}$is minimized subject to the constraint of Equation (8a).

The set of weights, {w_(i) ^(D)} and {w_(i) ^(U)}, and the sets ofspatial signatures, {a_(i) ^(D)} and {a_(i) ^(U)}, of existingconnections are usually known or can be determined by known methods bythe base station(s). See for example, above mentionedincorporated-by-reference U.S. Pat. No. 5,592,490. Any means for sodetermining these quantities is called a spatial processor herein.

The path gains, {L_(i) ^(D)} and {L_(i) ^(U)}, can be estimated asfollows. First, the RSSI is estimated as E(R²)≈ R² according to Equation(19b) as described below in the description of the “signal qualityestimation” aspect of the invention. The RSSI is then measured after thereceive spatial demultiplexing for each spatial channel. Note that inthe preferred embodiment, the RSSI is available at input to thedemodulator, a decision feedback demodulator. See above-mentionedincorporated-by-reference U.S. patent application Ser. No. 08/729,390.This is used as before to estimate the signal quality as expressed bySINR. That is, $\begin{matrix}{L_{i}^{D} = {{RSSI}_{i}^{D}/{p_{i}^{D}\left( {1 + \frac{1}{\overset{\_}{\overset{\_}{{SINR}_{i}^{D}}}}} \right)}}} & \left( {9a} \right) \\{{L_{i}^{U} = {{RSSI}_{i}^{U}/{p_{i}^{U}\left( {1 + \frac{1}{\overset{\_}{\overset{\_}{{SINR}_{i}^{U}}}}} \right)}}},} & \left( {9b} \right)\end{matrix}$where SINR_(i) ^(D) and SINR_(i) ^(U) respectively are the downlink anduplink estimated signal to interference-plus-noise ratio experienced bythe ith subscriber unit and its associated base station (using themethods previously described as in Equation (19) and (20) and FIGS. 5and 6), and the p_(i) ^(D) and p_(i) ^(U) are the known downlink anduplink transmit power used by the base station associated with the ithsubscriber and base station associated with the ith subscriber duringeach of the last transmitted bursts.

As a first alternative to using Equations (9a) and (9b), since one knowsthe power being transmitted during initial call set up, the path losscan be obtained by measuring the average power at the antennas.

As another alternative, in some systems, such as systems using the IS-95CDMA standard, a pilot tone exists and is continually transmitted at aknown power level on the downlink. Using a pilot tone such in this waycan be used to determine the path loss. Thus several methods may be usedto determine the pass loss.

While the method of the invention is not restricted to such systems,note that for TDD systems such as PHS, it is reasonable to assume thatthe path loss is identical for the uplink and downlink.

The intercell downlink and uplink interference-plus-noise (I_(i) ^(D)and I_(i) ^(U)) are estimated as follows. Let K represent the timeupdate index and K−1 represent the previous time period. Then for thepresent (Kth) update, $\begin{matrix}{{I_{i}^{D}(K)} = {{{RSSI}_{i}^{D}(K)} - {{L_{i}^{D}(K)}{\sum\limits_{j = 1}^{d}{{{{w_{j}^{D^{*}}(K)}{a_{i}^{D}(K)}}}^{2}{p_{j}^{D}\left( {K - 1} \right)}}}}}} & \left( {10a} \right) \\{{I_{i}^{U}(K)} = {{{RSSI}_{i}^{U}(K)} - {\sum\limits_{j = 1}^{d}{{{{w_{i}^{U^{*}}(K)}{a_{j}^{U}(K)}}}^{2}{p_{j}^{U}\left( {K - 1} \right)}{{L_{j}^{U}(K)}.}}}}} & \left( {10b} \right)\end{matrix}$

In another alternate embodiment, the following formulation is used forestimating the downlink and uplink interference-plus-noise I_(i) ^(D)and I_(i) ^(U)). Again, K is used in the following equation as an indexto represent the value for the present calculation: $\begin{matrix}{{I_{i}^{D}(K)} = {\frac{{RSSI}_{i}^{D}(K)}{1 + \overset{\_}{\overset{\_}{{SINR}_{i}^{D}(K)}}} - {{L_{i}^{D}(K)}{\sum\limits_{j \neq i}{{{{w_{j}^{D^{*}}(K)}{a_{i}^{D}(K)}}}^{2}{p_{j}^{D}\left( {K - 1} \right)}}}}}} & \left( {11a} \right) \\{{I_{i}^{U}(K)} = {\frac{{RSSI}_{i}^{U}(K)}{1 + \overset{\_}{\overset{\_}{{SINR}_{i}^{U}(K)}}} - {\sum\limits_{j \neq i}{{{{w_{i}^{U^{*}}(K)}{a_{j}^{U}(K)}}}^{2}{p_{j}^{U}\left( {K - 1} \right)}{{L_{j}^{U}(K)}.}}}}} & \left( {11b} \right)\end{matrix}$

In one embodiment of explicitly solving the localized power controloptimization problem, Equation (7) (as part of downlink step 733) andEquation (8) (as part of uplink step 711) are each solved at each basestation as a linear programming problem. Any known method of solving thelinear programming problem may be used.

In a second (the preferred) embodiment of explicitly solving for thelocalized power controls, the constraints in Equations (7a) and (8b) aremodified to be equality constraints. That is, in the preferredembodiment, one determines the powers in the downlink and uplink,respectively, by solving Equations (7a) and (8a), respectively, withequality constraints. That is, in a conventional downlink channel, theset of powers to apply for downlink communications for the spatialchannels in any conventional downlink channel are determined be settingthe predicted downlink SINR in each downlink spatial channel of theconventional downlink channel to be equal to a target SINR for thatdownlink spatial channel. In the preferred embodiment, the target SINRis the same for all downlink spatial channels of the conventional uplinkchannel. Also, in a conventional uplink channel, the set of powers toapply for uplink communications for the spatial channels in anyconventional uplink channel are determined be setting the predicteduplink SINR in each uplink spatial channel of the conventional uplinkchannel to be equal to a target SINR for that uplink spatial channel. Inthe preferred embodiment, the target SINR is the same for all uplinkspatial channels of the conventional uplink channel.

Assuming that solving the localized power control minimization problemat each base station eventually leads to a globally optimal solution(when no quantities such as path gains change, there are no new calls,etc.), using equality is physically intuitive: in order to minimizetotal power, one should only allocate enough power to satisfy thedesired acceptable level of performance, and no more.

When Equations (7a) and (8a) are modified to be equality constraints,each of these equations may then be transformed to linear equations ofthe form, for use in downlink step 733, $\begin{matrix}{{{L_{i}^{D}{{w_{i}^{D^{*}}a_{i}^{D}}}^{2}p_{i}^{D}} - {{SINR}_{{target}_{i}}^{2}{\sum\limits_{{j \neq i},{j = 1}}^{d}{L_{i}^{D}{{w_{i}^{D^{*}}a_{i}}}^{2}p_{j}^{D}}}}} = {I_{i}^{D}{SINR}_{{target}_{i}}^{D}}} & \left( {12a} \right)\end{matrix}$and for use in uplink step 711, $\begin{matrix}{{{L_{i}^{U}{{w_{i}^{U^{*}}a_{i}^{U}}}^{2}} - {{SINR}_{{target}_{i}}^{U}{\sum\limits_{{j \neq i},{j = 1}}^{d}{L_{j}^{U}{{w_{i}^{U^{*}}a_{j}}}^{2}p_{j}^{U}}}}} = {I_{i}^{U}{{SINR}_{{target}_{i}}^{U}.}}} & \left( {12b} \right)\end{matrix}$Each of these will immediately be recognized to be a set of linearequations expressible in matrix form asAp=bwhere A is a square matrix of dimension d (the number of spatialchannels), p is the vector of powers (the superscripts U and D are leftout for simplicity), and b is a vector of target SINRs multiplied by theinterference and noise (I_(i)) quantities. Sets of Equations (12a) or(12b), when expressed in matrix form, each have an exact solution in thematrix formp=A ⁻¹ b.  (13)Substituting the values of L_(i), w_(i), a_(i), I_(i), and SINR_(i)(again, the superscripts U and D are left out for simplicity) producesthe values of the p_(i) exactly.

If any of the so obtained p_(i) values is negative, there is no feasiblesolution to the optimal power control problem. Several options exist. Inthe preferred embodiment, channel reassignment is carried out accordingto the channel reassignment procedure for the system.

FIGS. 8(a) and 8(b), respectively, are flow diagrams showing using thissecond embodiment for steps 711 and 733, respectively, for uplink anddownlink ongoing power control, respectively. FIG. 8(a) shows such anembodiment 801 of step 711 for applying on-going uplink power controlfor a given SDMA channel i. In step 803, the process obtains from thespatial processor, for all i, the uplink quantities {w_(i) ^(U)} and{a_(i) ^(U)}. Then, in step 805, for all i, using the estimates SINR_(i)^(U) obtained (step 709), the sets of path gains, {L_(i) ^(U)}, arecomputed using Equations (9b), and the interference plus noisequantities {I_(i) ^(U)} also are determined using Equations (10b) or(11b). In step 807, the uplink power assignments are determined,preferably by solving the equality constraint problems (Equations (12b)and (13)). In step 809, the uplink power levels are adjusted inaccordance with the solution obtained in step 807 by commanding the SUsto use such powers, and the process ends (returning to step 704). FIG.8(b) similarly shows an embodiment 821 of step 721 for applying on-goingdownlink power control for a given SDMA channel i. In step 823, theprocess obtains from the spatial processor, for all i, the downlinkquantities {w_(i) ^(D)} and {a_(i) ^(D)}. Then, in step 825, for all i,the sets of path gains, {L_(i) ^(D)}, are computed using Equations (9a),and the interference plus noise quantities {I_(i) ^(D)} also aredetermined using Equations (10a) or (11a). In step 827, the the downlinkpower assignments are determined, preferably by solving the equalityconstraint problems (Equations (12a) and (13)). In step 829, thedownlink power levels are adjusted in accordance with the solutionobtained in step 827, and the process ends (returning to step 724).

Signal Quality Estimation

Step 313 of method 301 for initial power control included determiningthe signal quality. Also, both the ongoing power control method usingEquations (3) or (4) and the localized power control method explicitlysolving Equations (7) and (8) include using an estimate of a measure(the SINR) of the signal quality (see steps 709 and 729 in flow charts701 and 721, respectively). Another aspect of the invention is a methodfor implementing such steps. While any methods for determining theseestimates may be used in implementing the power control aspects of theinvention, another aspect of the invention is a RF carrier signalquality estimator method and apparatus which is applicable to allangle-modulated RF carriers. Because of the large variety of theseangle-modulated systems, the detailed description will be only for oneof two types in order to increase understanding of the invention. Thetwo sample angle-modulated signals selected for this purpose are aquaternary phase shift keyed (QPSK) signal and a differential quaternaryphase shift keyed signal (DQPSK) that are found in wide-spread use inthe communication field. Each symbol in these schemes contains two bits(a dibit) of information. The important feature of the phase modulatedsignal with respect to the signal quality estimation aspect of thecurrent invention is that the magnitude of the data symbols is assumedto be constant in the absence of noise or other forms of corruption. Thenumber of discrete phase levels (four in these cases) is not important.The difference between these two signals is that in DQPSK, a dibit ismapped onto the phase difference between two successive symbols, whilein QPSK, a dibit is mapped onto the phase of the symbol itself. Thus,the phase plane of a QPSK signal is the same as the differential phaseplane of a QPSK signal, which is the phase plane of the difference inphase between two successive symbols. Also, in QPSK, the four symbolpoints used in the description are 0, π2, π, and 3π/2, while in theparticular DQPSK used in the preferred embodiment, π/4 DQPSK, the foursymbol points used are, in the differential phase plane, π/4, 3π/4, 5π/4and 7π/4 (that is, ±π/4 and ±3π/4). That is, the above QPSK phase planeis rotated by π/4 for the π/4 DQPSK case. It will be clear to those ofordinary skill in the art how to implement for one case from adescription of the other case. From the following description, it willalso be clear to those of ordinary skill in the art how to adapt theapplication of the principles to other forms of angle-modulationsystems.

FIG. 4 is a complex phase plane diagram 401 of the four states of a QPSKmodel, except that an extraneous noise vector, ΔS, and resultant phaseerror, Δθ, have been added to represent the practical situation in whichthe presence of noise and interference introduces both amplitude andphase errors into the observed data symbol R. Vector ΔS corresponds tothe vector difference between the uncontaminated symbol S and theobserved symbol R.

FIG. 4 is a complex phase plane representation of a received data symbolsignal R on the phase plane. Also shown on the phase plane 401 are thefour decision points 403, 404, 405, and 406 at phases 0, π/2, π, and3π/2, respectively (the constellation of decisions). Any frequencyoffsets present may be thought of as rotations of the constellationpoints relative to the received signal R. R may be thought of as aconstant modulus signal S, onto which an extraneous noise vector ΔS hasbeen added, resulting in phase error Δθ. ΔS represents noise andinterference which in practice is present and introduces both amplitudeand phase errors into the observed data symbol R. That is, vector ΔScorresponds to the vector difference between the uncontaminated symbol Sand the observed symbol R. The SINR thus may be estimated by estimatingthe ratio of the squared magnitudes of S and ΔS. That is,SINR=E[S²]/E[ΔS²]. The essence of the preferred embodiment signalquality estimator is to estimate the SINR of the received signalentirely from observations of radius R over a burst of received data.This exploits the fact that radius R is invariant to rotations of theconstellation, thus making the SINR estimate substantially immune tofrequency offsets.

For the purpose of explanation, the noise and interference vector, ΔS,is modeled as a zero-mean Gaussian random process with independent real(i.e., in phase I) and imaginary (i.e., quadrature Q) components, eachcomponent with a variance of σ². This model is chosen because it leadsto a realistic and practical method for determining the modulus S andthe statistical evaluation of the associated noise and interferencevector. While this model is used in explaining the method, the methodworks for actual signals and noise for which the zero-mean Gaussianrandom process assumption may not hold.

An embodiment is now described of a method for estimating thesignal-to-interference-plus-noise (SINR) for a received signal. This isapplicable for power control applications, including step 313 of powercontrol method 301 and in each of the steps in the ongoing power controlembodiments that include having an SINR estimate. In FIG. 4, vector ΔSmay be represented as the sum of two orthogonal zero-mean Gaussian noisecomponents, one noise component (n₁) in-phase with carrier vector S, andthe other a quadrature component (n₂), both n₁ and n₂ each having avariance of σ², as shown in FIG. 4. The quantity to be estimated, theSINR, is thenSINR=E[S ²]/2σ².  (14)

If the modulus of signal S is substantially constant over a burst, thenreceived signal amplitude R is approximately Rician distributed, with$\begin{matrix}{{{{E\lbrack R\rbrack} = {\sqrt{2\sigma^{2}}{f({SINR})}}},{where}}{{{f({SINR})} = {{\mathbb{e}}^{- {SINR}}{\sum\limits_{l = 0}^{\infty}\frac{{\Gamma\left( {\frac{3}{2} + l} \right)}{SINR}^{l}}{{\Gamma\left( {\frac{1}{2} + l} \right)}{l!}}}}},{and}}} & (15) \\\begin{matrix}{{E\left\lbrack R^{2} \right\rbrack} = {{E\left\lbrack S^{2} \right\rbrack} + {2\sigma^{2}}}} \\{= {2{{\sigma^{2}\left( {1 + {SINR}} \right)}.}}}\end{matrix} & (16)\end{matrix}$

Given values of E[R] and E[R²], Equations (15) and (16) are two(nonlinear) equations in two unknowns (σ² and SINR). Thus one embodimentof the method is to simultaneously solve Equations (15) and (16) forSINR. Values of ƒ(SINR) according may be pre-stored in a lookup table.Alternatively, the approximation ƒ(SINR)≈1 may be used. In the preferredembodiment, the complex valued baseband signal for the communicationschannel preferably is provided as in-phase and quadrature components,denoted by l and Q, respectively. The values of E{R^(k)}, k=1, 2, etc.,are estimated by determining R²=(I²+Q²) for each sample in a burst. Notethat the average of the (I²+Q²) values over a set of samples is ameasure of the received signal strength indicator (RSSI), commonlyavailable in receivers. Denote the sampled values of I and Q by I(n) andQ(n), respectively, where each successive sample n is ideally an on-baudsampling point, the on-baud sampling point for a given pulse-shapedsymbol corresponds to the center point in time of the pulse-shapedsymbol. In practice, due to imperfections in timing alignment, theon-baud sampling point corresponds to the time sample closest to thecenter of the pulse-shaped symbol. Thus, successive samples in I(n) andQ(n) are one baud period apart. Note that these I and Q values are for asingle modulated signal of a single (spatial) channel. Thus, in thesystem of the preferred embodiment, these are the I and Q values afterspatial processing and substantially on the baud points. In the receiverof the transceiver of FIG. 1, the output downconverter/filter 131 arethe sampled signals for one antenna only—that is, before spatialprocessing, and are oversampled. Determining signals I(n) and Q(n) afterspatial processing at or close to the baud points is thus assumed tohave been carried out, preferably in the corresponding RX DSP 209. SeeOur Demodulation Patent (above mentioned U.S. patent application Ser.No. 08/729,390) for a discussion of one example of processing includingdetermining signals I(n) and Q(n) after spatial processing at or closeto the baud points. Denote by N the number of samples in a burst. Thesampled modulus information is extracted by forming the sum of thesquares of the in phase and quadrature signals,R(n)=√{square root over (I ²(n)+Q ²(n))},  (17a)R ²(n)=I ²(n)+Q ²(n), and  (17b)R ⁴(n)=(I ²(n)+Q ²(n))².  (17c)

E{R^(k)} is then approximated by computing the ensemble average, denotedR^(k) , over the burst, $\begin{matrix}{{\overset{\_}{R^{k}} = {\frac{1}{N}{\sum\limits_{i = 1}^{N}{R^{k}(n)}}}},} & (18)\end{matrix}$which, for the cases k=1, 2 and 4, $\begin{matrix}{\overset{\_}{R} = {\frac{1}{N}{\sum\limits_{i = 1}^{N}\sqrt{\left( {{I^{2}(n)} + {Q^{2}(n)}} \right)}}}} & \left( {19a} \right) \\{{\overset{\_}{R^{2}} = {{\frac{1}{N}{\sum\limits_{i = 1}^{N}{I^{2}(n)}}} + {Q^{2}(n)}}},{and}} & \left( {19b} \right) \\{\overset{\_}{R^{4}} = {\frac{1}{N}{\sum\limits_{i = 1}^{N}{\left( {{I^{2}(n)} + {Q^{2}(n)}} \right)^{2}.}}}} & \left( {19c} \right)\end{matrix}$

R² then can be used as a measure of the RSSI.

In one embodiment, an iterative solution is used in which values of 2σ²and SINR are assumed and substituted into Equation (15) and (16), thecomputed value of E{R²} and E{R} are compared with the estimated RSSI (R² obtained from measurement) and R, respectively, and, if thedifferences are acceptably small, accepting the substituted values asthe solutions that lead to the SINR estimate. This is a standarditerative way of solving the two simultaneous nonlinear Equations (15)and (16).

Note that in the above, in order to estimate E{R} from measurements,(see Equation (17a)), a square root operation needs to be performed.This is computationally expensive.

A second preferred method is based only on using the even powerednon-central moments, E{R²} and E{R⁴}, determined according to Equations(17b) and (17c). The even powered non-central moments, E{R²} and E{R⁴}are the mean power and the mean squared power measurements. Note thatonce the instantaneous power R²(n)=I²(n)+Q²(n) is determined (for use inEquations (17b)), determining the squared power R⁴(n)=[R²(n)]² requiresonly a single additional multiplication per sample, and the estimatedsignal-to-noise ratio is determined, preferably with at most one squareroot operation, using $\begin{matrix}\begin{matrix}{{SINR} = \frac{\sqrt{2 - \frac{\overset{\_}{R^{4}}}{\left( \overset{\_}{R^{2}} \right)^{2}}}\quad}{1 - \sqrt{2 - \frac{\overset{\_}{R^{4}}}{\left( \overset{\_}{R^{2}} \right)^{2}}}}} \\{{= \frac{A + \sqrt{A}}{1 - A}},{{{where}\quad A} = {2 - {\frac{\overset{\_}{R^{4}}}{\left( \overset{\_}{R^{2}} \right)^{2}}.}}}}\end{matrix} & (20)\end{matrix}$

FIG. 5(a) is a block diagram of one realization of a base stationapparatus (501) used for obtaining the signal quality estimates. Thisapparatus is a simplified version of the base station apparatus of FIGS.1 and 2. The apparatus 501 is comprised of an antenna array 503 forreceiving RF signals, a set of RF receivers 505 (i.e., antenna receiveapparatuses) for converting the signals at each of the elements inantenna 503 into a complex valued baseband signals, a spatial processor507 for determining the baseband signals from a particular remote userpreferably provided as in-phase (I) component 509 and quadrature (Q)component 511, signals 509 and 511 determined substantially at the baudpoints. The I and Q signals 509 and 511, respectively, are fed to asignal quality estimator 513 for further processing to produce thedesired signal quality estimate, shown as signal quality indicator 515in FIG. 5(a). In the base station of the preferred embodiment, eachelement of antenna array 503 and each RF receiver 505 are implemented inthe transceiver module shown in FIG. 1, and generate digital pre-spatialprocessing signals as the outputs of downconverter/filters 131. Spatialprocessor 507 is preferably a programmable digital arithmetic processor.When used in the base station of the preferred embodiment, spatialprocessor 507 is part of the modem module of FIG. 2, in particular, oneof the RX DSPs 209, the particular RX DSP being the one for the slotbeing received. Signal quality processor 513 also is preferably aprogrammable digital arithmetic processor. When used in the base stationof the preferred embodiment, signal quality processor 513 is part of themodem module of FIG. 2, in particular, one of the RX DSPs 209, theparticular RX DSP being the one for the slot being received.

FIG. 5(b) is a block diagram of one realization of a subscriber unitapparatus (521) used for obtaining the signal quality estimates. Thisapparatus is a simplified version of the subscriber unit apparatus ofFIGS. 9 and 10. The apparatus 521 is comprised of an antenna 523 forreceiving RF signals, an RF receiver 525 for converting the signal fromantenna 523 into a complex valued baseband signal, preferably asin-phase (I) component 529 and quadrature (Q) component 531, signals 529and 531 determined substantially at the baud points. The I and Q signals529 and 531, respectively, are fed to a signal quality estimator 533 forfurther processing to produce the desired signal quality estimate, shownas signal quality indicator 535 in FIG. 5(b). In the subscriber unit ofthe preferred embodiment, antenna 523 and RF receiver 525 areimplemented in the RF system shown in FIG. 9, and generate oversampled Iand Q signals. Signal quality processor 533 also is preferably aprogrammable digital arithmetic processor. When used in the subscriberunit of the preferred embodiment, signal quality processor 533 isDSP(RX) 1042, and its function includes determining the samples at theapproximate baud points.

In the case of digital processing, signals 509 and 511 (in a BS) andsignals 529 and 531 (after baud point processing by DSP(RX) 1042 in theSU) are sampled at the estimated baud points. These signals as beforeare denoted by I(n) and Q(n), respectively, where each successive samplen is at or close to successive baud points. The sampled modulusinformation is extracted by forming the sum of the squares of the inphase and quadrature signals, that is, the instantaneous power. Thus,instantaneous power R²(n) is obtained using Equation (17b), andinstantaneous squared power R⁴(n) is obtained as [R²(n)]²=R²(n)*R²(n).

The second and fourth moments, means power R² and mean squared power R⁴, respectively, are estimated by forming the averages of R²(n) and R⁴(n)over a moving window. Such moving averages may be determined usingEquations (19b) and (19c) for the cases k=2 and k=4.

Equation (20) is now used to determine the quality estimate. Inpractice, as would be clear to one of ordinary skill in the art, onlythe summations in Equation (19b) (and Equation (19c), if used) areformed. The 1/N factor need not be determined for all averages so longas the correct scaling is maintained in the determination of the qualityestimate using Equation (20).

In an improved embodiment, rather than determine the SINR over a singleburst, an additional step is added of taking a moving average of SINRsover several bursts. For example, the Kth running average determined,after K bursts, asSINR_(K)=αSINR+(1−α)SINR_(K−1)  (21)where 0<α<1, SINR is the new measure for the current (Kth) burst, andSINR_(K) denotes the Kth running average of the SINR. The value of α isselected to control the rate of adaptation of the moving average tochanging conditions. In the preferred embodiment of a PHS system, avalue of 0.8 is used for α. As would be clear to one of ordinary skillin the art, the running average of Equation (21) is easily implementedas a finite impulse response (FIR) filter, and does not add much to thecomputational burden, as the data rate (a new SINR every burst) is low.The moving average preferably is implemented in signal quality processor513 (in a base station) and processor 533 (in a SU).

FIG. 6 is a flow diagram for method 601 that summarizes the method forobtaining a signal quality estimate in angle modulated communicationsystems. In step 603, an angle-modulated signal is received. Step 605generates the in-phase (I) and quadrature (Q) components of the receivedsignal in baseband, after spatial processing in the case of SDMA, andsubstantially at the baud points in the case of digital processing. Step607 extracts estimates of at least two distinct moments of the modulusof the received signal from the I, Q components, and step 609 determinesaverages of the moments. Step 611 determines the signal quality estimateas the SINR estimate.

The methods and apparatuses described herein for controlling transmitterpower level were, for sake of clarity, limited to specific cellularcommunication systems and implementations. For those of ordinary skillin the art, the application of the invention to other communicationsystems, such systems that use other air interfaces, systems that aredesigned for data transmission, analog systems, wireless local areanetworks (LANs), etc., will become evident from the description providedwithout departing from the spirit and scope of the invention whichshould only be limited as set forward in the claims that follow. Also,the specific methods and apparatuses described for estimating thequality of a received angle-modulated RF carrier was by way of exampleonly and should not be limited except as set forward in the claims thatfollow.

1. A method comprising: receiving, at a second radio station, a firstradio signal from a first radio station, the first signal beingtransmitted with a first assigned power level and a first transmitspatial weight vector; determining a quality of the received firstsignal; receiving a second radio signal from the first station, thesecond signal being transmitted using a second assigned power level anda second transmit spatial weight vector, the second assigned power levelbeing determined based on the determined quality of the received firstsignal; and determining the quality of the received second signal. 2.The method of claim 1, further comprising communicating the determinedquality of the first and second received radio signals to the firstradio station.
 3. The method of claim 1, further comprising determiningthe second power assignment at the second radio based on the determinedquality of the first received radio signal and communicating the secondpower assignment to the first station before receiving the secondsignal.
 4. The method of claim 1, further comprising: determining thesecond transmit spatial weight vector at the second station based on thedetermined quality of the first signal; communicating the secondtransmit spatial weight vector to the first station; and receiving thesecond signal from the first station using the second power assignmentand the second transmit spatial weight vector.
 5. The method of claim 4,further comprising repeating at least a portion of: receiving a signalfrom the first station; determining a quality of the signal; determininga power assignment based on the determined quality; determining atransmit spatial weight vector based on the determined quality;communicating the second power assignment to the first station;communicating the second transmit weight vector to the first station;and receiving a second signal from the first station using the secondpower assignment and the second transmit spatial weight vector.
 6. Amachine-readable medium comprising data that when operated on by themachine cause the machine to perform operations comprising: receiving atsecond radio station a first radio signal from a first radio station,the first signal being transmitted with a first assigned power level anda first transmit spatial weight vector; determining a quality of thereceived first signal; receiving a second radio signal from the firstradio station, the second signal being transmitted using a secondassigned power level and a second transmit spatial weight vector, thesecond assigned power level being determined based on the determinedquality of the received first signal; and determining the quality of thereceived second signal.
 7. The medium of claim 6, the operations furthercomprising communicating the determined quality of the first and secondreceived radio signals to the first radio station.
 8. The medium ofclaim 6, the operations further comprising repeating at least a portionof: receiving a signal from the first station; determining a quality ofthe signal; determining a power assignment based on the determinedquality; determining a transmit spatial weight vector based on thedetermined quality; communicating the second power assignment to thefirst station; communicating the second transmit weight vector to thefirst station; and receiving a second signal from the first stationusing the second power assignment and the second transmit spatial weightvector.
 9. A radio station comprising: a receiver to receive a firstradio signal from a second radio station, the first signal beingtransmitted with a first assigned power level and a first transmitspatial weight vector; a signal processor to determine a quality of thereceived first signal; the receiver further to receive a second radiosignal from the first radio station, the second signal being transmittedusing a second assigned power level and a second transmit spatial weightvector, the second assigned power level being determined based on thedetermined quality of the received first signal; and the signalprocessor further to determine the quality of the received secondsignal.
 10. The radio station of claim 9, wherein the signal processorfurther determines the second transmit spatial weight vector based onthe determined quality of the first signal, further comprising atransmitter to communicate the second transmit spatial weight vector tothe first station, and wherein the receiver further receives the secondsignal from the first station using the second power assignment and thesecond transmit spatial weight vector.
 11. The radio station of claim 9,wherein the receiver receives on a downlink channel and the transmittertransmits on an uplink channel.
 12. The radio station of claim 9,wherein the radio station comprises a component of a mobile cellulartelephone.
 13. A method comprising: transmitting a first radio signalfrom a first radio station to a second radio station, the first signalbeing sent according to a first power assignment and a first spatialweight vector; receiving from the second station a quality of the firstsignal; determining at the first station a second power assignment basedon the received quality of the first signal; and transmitting a secondradio signal from the first station to the second station using thesecond power assignment.
 14. The method of claim 13, further comprising:determining a second spatial weight vector at the first station based onthe received quality of the first signal; and wherein transmitting thesecond signal comprises transmitting the second signal using the secondpower assignment and the second spatial weight vector.
 15. The method ofclaim 13, further comprising: setting up an initial power assignment andspatial weight vector as the first power assignment and the firstspatial weight vector, the initial assignments being the same for allradios to which the first station transmits a first signal.
 16. Themethod of claim 13, wherein the second power assignment is applied as ascaling factor to the second spatial weight vector.
 17. The method ofclaim 13, wherein receiving from the second station comprises receivingan uplink signal.
 18. A machine-readable medium comprising data thatwhen operated on by the machine cause the machine to perform operationscomprising: transmitting a first radio signal from a first radio stationto a second radio station, the first signal being sent according to afirst power assignment and a first spatial weight vector; receiving fromthe second radio station a quality of the first signal; determining asecond power assignment based on the received quality of the firstsignal; and transmitting a second radio signal from the first radiostation to the second radio station using the second power assignment.19. The medium of claim 18, the operations further comprising:determining a second spatial weight vector at the first station based onthe received quality of the first signal; and wherein transmitting thesecond signal comprises transmitting the second signal using the secondpower assignment and the second spatial weight vector.
 20. The medium ofclaim 18, the operations further comprising: repeating at least aportion of: transmitting an additional radio signal to the secondstation; receiving a quality of the additional signal from the secondstation; determining an additional power assignment based on thereceived quality; determining an additional spatial weight vector basedon the received quality; and transmitting an additional signal to thesecond station using the determined additional power assignment and thedetermined additional spatial weight vector.
 21. A radio stationcomprising: a transmitter to transmit a first radio signal from theradio station to a second radio station, the first signal being sentaccording to a first power assignment and a first spatial weight vector;a receiver to receive from the second radio station a quality of thefirst signal; a signal processor to determine a second power assignmentbased on the received quality of the first signal; and the transmitterfurther to transmit a second radio signal from the first radio stationto the second radio station using the second power assignment.
 22. Theradio station of claim 21, wherein the signal processor is further todetermine a second spatial weight vector based on the received qualityof the first signal, and wherein the transmitter is further to transmitthe second signal using the second power assignment and the secondspatial weight vector.
 23. The radio station of claim 21, wherein thetransmitter transmits to a plurality of second radio stations as a basestation in cellular radio communication system and wherein the signalprocessor further sets up an initial power assignment and spatial weightvector as the first power assignment and the first spatial weightvector, the initial assignments being the same for all radios to whichthe transmitter transmits a first signal.
 24. The radio station of claim21, wherein the signal processor applies the second power assignment asa scaling factor to the second spatial weight vector.